Radio with antenna array and multiple RF bands

ABSTRACT

A intelligent backhaul radio is disclosed that is compact, light and low power for street level mounting, operates at 100 Mb/s or higher at ranges of 300 m or longer in obstructed LOS conditions with low latencies of 5 ms or less, can support PTP and PMP topologies, uses radio spectrum resources efficiently and does not require precise physical antenna alignment.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a Continuation-in-part of U.S. patent applicationSer. No. 15/403,713, filed on Jan. 11, 2015, which is a Continuation ofU.S. patent application Ser. No. 15/203,658, filed on Jul. 6, 2016, nowU.S. Pat. No. 9,578,643, which is a Continuation of U.S. patentapplication Ser. No. 14/988,578, filed on Jan. 5, 2016, now U.S. Pat.No. 9,408,215, which is a Continuation of U.S. patent application Ser.No. 14/686,674, filed on Apr. 14, 2015, now U.S. Pat. No. 9,282,560,which is a Continuation of U.S. patent application Ser. No. 14/337,744,filed on Jul. 22, 2014, now U.S. Pat. No. 9,055,463, which is aContinuation of U.S. patent application Ser. No. 13/645,472, filed onOct. 4, 2012, now U.S. Pat. No. 8,811,365, which is a Continuation ofU.S. patent application Ser. No. 13/371,366, filed on Feb. 10, 2012, nowU.S. Pat. No. 8,311,023, which is a Continuation of U.S. patentapplication Ser. No. 13/212,036, filed on Aug. 17, 2011, now U.S. Pat.No. 8,238,318, the disclosures of which are hereby incorporated hereinby reference in their entireties.

The present application is also related to U.S. Provisional PatentApplication No. 61/857,661, filed on Jul. 23, 2013, and U.S. patentapplication Ser. No. 14/151,190, filed on Jan. 9, 2014, now U.S. Pat.No. 8,982,772, U.S. patent application Ser. No. 14/197,158, filed onMar. 4, 2014, now U.S. Pat. No. 8,928,542, U.S. patent application Ser.No. 14/199,734, filed on Mar. 6, 2014, now U.S. Pat. No. 8,872,715, U.S.patent application Ser. No. 14/559,859, filed on Dec. 3, 2014, U.S.patent application Ser. No. 13/536,927, filed on Jun. 28, 2012, now U.S.Pat. No. 8,467,363, U.S. patent application Ser. No. 13/898,429, filedon May 20, 2013, now U.S. Pat. No. 8,824,442, U.S. patent applicationSer. No. 14/336,958, filed on Jul. 21, 2014, now U.S. Pat. No.9,001,809, U.S. patent application Ser. No. 14/632,624, filed on Feb.26, 2015, now U.S. Pat. No. 9,178,558, U.S. patent application Ser. No.14/837,797, filed on Aug. 26, 2015, now U.S. Pat. No. 9,350,411, U.S.patent application Ser. No. 15/142,793, filed Apr. 29, 2016, U.S. patentapplication Ser. No. 14/608,024, filed on Jan. 28, 2015, now U.S. Pat.No. 9,345,036, U.S. patent application Ser. No. 15/050,009, filed onFeb. 22, 2016, U.S. Provisional Application No. 62/130,100, filed Mar.9, 2015, U.S. Provisional Application No. 62/135,573, filed Mar. 19,2015, U.S. Provisional Application No. 61/910,194, filed Nov. 29, 2013,U.S. patent application Ser. No. 14/498,959, filed on Sep. 26, 2014, nowU.S. Pat. No. 9,049,611, U.S. patent application Ser. No. 14/688,550,filed on Apr. 16, 2015, now U.S. Pat. No. 9,313,674, U.S. patentapplication Ser. No. 15/060,013, filed on Mar. 3, 2016, U.S. patentapplication Ser. No. 14/098,456, filed on Dec. 5, 2013, now U.S. Pat.No. 8,989,762, U.S. patent application Ser. No. 14/502,471, filed Sep.30, 2014, U.S. patent application Ser. No. 14/624,365, filed on Feb. 17,2015, U.S. patent application Ser. No. 14/666,294, filed Mar. 23, 2015,U.S. patent application Ser. No. 13/448,294, filed on Apr. 16, 2012, nowU.S. Pat. No. 8,385,305, U.S. patent application Ser. No. 13/748,544,filed on Jan. 23, 2013, now U.S. Pat. No. 8,942,216, U.S. patentapplication Ser. No. 14/552,431, filed on Nov. 24, 2014, now U.S. Pat.No. 9,226,295, U.S. patent application Ser. No. 14/950,354, filed onNov. 24, 2015, now U.S. Pat. No. 9,374,822, U.S. patent application Ser.No. 15/165,504, filed on May 26, 2016, U.S. patent application Ser. No.13/271,051, filed on Oct. 11, 2011, now U.S. Pat. No. 8,761,100, U.S.patent application Ser. No. 13/415,778, filed on Mar. 8, 2012, now U.S.Pat. No. 8,300,590, U.S. patent application Ser. No. 13/632,961, filedon Oct. 1, 2012, U.S. patent application Ser. No. 13/632,993, filed onOct. 1, 2012, now U.S. Pat. No. 9,226,315, U.S. patent application Ser.No. 13/633,028, filed on Oct. 1, 2012, now U.S. Pat. No. 8,830,943, U.S.patent application Ser. No. 14/964,292, filed on Dec. 9, 2015, U.S.patent application Ser. No. 13/371,346, filed on Feb. 10, 2012, now U.S.Pat. No. 8,502,733, U.S. patent application Ser. No. 13/934,175, filedon Jul. 2, 2013, now U.S. Pat. No. 9,179,240, U.S. patent applicationSer. No. 14/839,018, filed on Aug. 28, 2015, now U.S. Pat. No. 9,325,398and U.S. patent application Ser. No. 15/084,867, filed on Mar. 30, 2016,the disclosures of which are incorporated herein by reference in theirentireties.

BACKGROUND 1. Field

The present disclosure relates generally to data networking and inparticular to a backhaul radio for connecting remote edge accessnetworks to core networks.

2. Related Art

Data networking traffic has grown at approximately 100% per year forover 20 years and continues to grow at this pace. Only transport overoptical fiber has shown the ability to keep pace with thisever-increasing data networking demand for core data networks. Whiledeployment of optical fiber to an edge of the core data network would beadvantageous from a network performance perspective, it is oftenimpractical to connect all high bandwidth data networking points withoptical fiber at all times. Instead, connections to remote edge accessnetworks from core networks are often achieved with wireless radio,wireless infrared, and/or copper wireline technologies.

Radio, especially in the form of cellular or wireless local area network(WLAN) technologies, is particularly advantageous for supportingmobility of data networking devices. However, cellular base stations orWLAN access points inevitably become very high data bandwidth demandpoints that require continuous connectivity to an optical fiber corenetwork.

When data aggregation points, such as cellular base station sites, WLANaccess points, or other local area network (LAN) gateways, cannot bedirectly connected to a core optical fiber network, then an alternativeconnection, using, for example, wireless radio or copper wirelinetechnologies, must be used. Such connections are commonly referred to as“backhaul.”

Many cellular base stations deployed to date have used copper wirelinebackhaul technologies such as T1, E1, DSL, etc. when optical fiber isnot available at a given site. However, the recent generations of HSPA+and LTE cellular base stations have backhaul requirements of 100 Mb/s ormore, especially when multiple sectors and/or multiple mobile networkoperators per cell site are considered. WLAN access points commonly havesimilar data backhaul requirements. These backhaul requirements cannotbe practically satisfied at ranges of 300 m or more by existing copperwireline technologies. Even if LAN technologies such as Ethernet overmultiple dedicated twisted pair wiring or hybrid fiber/coax technologiessuch as cable modems are considered, it is impractical to backhaul atsuch data rates at these ranges (or at least without adding intermediaterepeater equipment). Moreover, to the extent that such special wiring(i.e., CAT 5/6 or coax) is not presently available at a remote edgeaccess network location; a new high capacity optical fiber isadvantageously installed instead of a new copper connection.

Rather than incur the large initial expense and time delay associatedwith bringing optical fiber to every new location, it has been common tobackhaul cell sites, WLAN hotspots, or LAN gateways from offices,campuses, etc. using microwave radios. An exemplary backhaul connectionusing the microwave radios 132 is shown in FIG. 1. Traditionally, suchmicrowave radios 132 for backhaul have been mounted on high towers 112(or high rooftops of multi-story buildings) as shown in FIG. 1, suchthat each microwave radio 132 has an unobstructed line of sight (LOS)136 to the other. These microwave radios 132 can have data rates of 100Mb/s or higher at unobstructed LOS ranges of 300 m or longer withlatencies of 5 ms or less (to minimize overall network latency).

Traditional microwave backhaul radios 132 operate in a Point to Point(PTP) configuration using a single “high gain” (typically >30 dBi oreven >40 dBi) antenna at each end of the link 136, such as, for example,antennas constructed using a parabolic dish. Such high gain antennasmitigate the effects of unwanted multipath self-interference or unwantedco-channel interference from other radio systems such that high datarates, long range and low latency can be achieved. These high gainantennas however have narrow radiation patterns.

Furthermore, high gain antennas in traditional microwave backhaul radios132 require very precise, and usually manual, physical alignment oftheir narrow radiation patterns in order to achieve such highperformance results. Such alignment is almost impossible to maintainover extended periods of time unless the two radios have a clearunobstructed line of sight (LOS) between them over the entire range ofseparation. Furthermore, such precise alignment makes it impractical forany one such microwave backhaul radio to communicate effectively withmultiple other radios simultaneously (i.e., a “point to multipoint”(PMP) configuration).

In wireless edge access applications, such as cellular or WLAN, advancedprotocols, modulation, encoding and spatial processing across multipleradio antennas have enabled increased data rates and ranges for numeroussimultaneous users compared to analogous systems deployed 5 or 10 yearsago for obstructed LOS propagation environments where multipath andco-channel interference were present. In such systems, “low gain”(usually <6 dBi) antennas are generally used at one or both ends of theradio link both to advantageously exploit multipath signals in theobstructed LOS environment and allow operation in different physicalorientations as would be encountered with mobile devices. Althoughimpressive performance results have been achieved for edge access, suchresults are generally inadequate for emerging backhaul requirements ofdata rates of 100 Mb/s or higher, ranges of 300 m or longer inobstructed LOS conditions, and latencies of 5 ms or less.

In particular, “street level” deployment of cellular base stations, WLANaccess points or LAN gateways (e.g., deployment at street lamps, trafficlights, sides or rooftops of single or low-multiple story buildings)suffers from problems because there are significant obstructions for LOSin urban environments (e.g., tall buildings, or any environments wheretall trees or uneven topography are present).

FIG. 1 illustrates edge access using conventional unobstructed LOS PTPmicrowave radios 132. The scenario depicted in FIG. 1 is common for many2^(nd) Generation (2G) and 3^(rd) Generation (3G) cellular networkdeployments using “macrocells”. In FIG. 1, a Cellular Base TransceiverStation (BTS) 104 is shown housed within a small building 108 adjacentto a large tower 112. The cellular antennas 116 that communicate withvarious cellular subscriber devices 120 are mounted on the towers 112.The PTP microwave radios 132 are mounted on the towers 112 and areconnected to the BTSs 104 via an nT1 interface. As shown in FIG. 1 byline 136, the radios 132 require unobstructed LOS.

The BTS on the right 104 a has either an nT1 copper interface or anoptical fiber interface 124 to connect the BTS 104 a to the Base StationController (BSC) 128. The BSC 128 either is part of or communicates withthe core network of the cellular network operator. The BTS on the left104 b is identical to the BTS on the right 104 a in FIG. 1 except thatthe BTS on the left 104 b has no local wireline nT1 (or optical fiberequivalent) so the nT1 interface is instead connected to a conventionalPTP microwave radio 132 with unobstructed LOS to the tower on the right112 a. The nT1 interfaces for both BTSs 104 a, 104 b can then bebackhauled to the BSC 128 as shown in FIG. 1.

FIG. 2 is a block diagram of the major subsystems of a conventional PTPmicrowave radio 200 for the case of Time-Division Duplex (TDD)operation, and FIG. 3 is a block diagram of the major subsystems of aconventional PTP microwave radio 300 for the case of Frequency-DivisionDuplex (FDD) operation.

As shown in FIG. 2 and FIG. 3, the conventional PTP microwave radiotraditionally uses one or more (i.e. up to “n”) T1 interfaces 204 (or inEurope, E1 interfaces). These interfaces 204 are common in remote accesssystems such as 2G cellular base stations or enterprise voice and/ordata switches or edge routers. The T1 interfaces are typicallymultiplexed and buffered in a bridge (e.g., the Interface Bridge 208,308) that interfaces with a Media Access Controller (MAC) 212, 312.

The MAC 212, 312 is generally denoted as such in reference to asub-layer of Layer 2 within the Open Systems Interconnect (OSI)reference model. Major functions performed by the MAC include theframing, scheduling, prioritizing (or “classifying”), encrypting anderror checking of data sent from one such radio at FIG. 2 or FIG. 3 toanother such radio. The data sent from one radio to another is generallyin a “user plane” if it originates at the T1 interface(s) or in the“control plane” if it originates internally such as from the Radio LinkController (RLC) 248, 348 shown in FIG. 2 or FIG. 3. A typical MAC frameformat 400 (known as a MAC protocol data unit, or “MPDU”) with header404, frame body 408 and frame check sum (FCS) 412 is shown in FIG. 4.

With reference to FIGS. 2 and 3, the Modem 216, 316 typically resideswithin the “baseband” portion of the Physical (PHY) layer 1 of the OSIreference model. In conventional PTP radios, the baseband PHY, depictedby Modem 216, 316, typically implements scrambling, forward errorcorrection encoding, and modulation mapping for a single RF carrier inthe transmit path. In receive, the modem typically performs the inverseoperations of demodulation mapping, decoding and descrambling. Themodulation mapping is conventionally Quadrature Amplitude Modulation(QAM) implemented with In-phase (I) and Quadrature-phase (Q) branches.

The Radio Frequency (RF) 220, 320 also resides within the PHY layer ofthe radio. In conventional PTP radios, the RF 220, 320 typicallyincludes a single transmit chain (Tx) 224, 324 that includes I and Qdigital to analog converters (DACs), a vector modulator, optionalupconverters, a programmable gain amplifier, one or more channelfilters, and one or more combinations of a local oscillator (LO) and afrequency synthesizer. Similarly, the RF 220, 320 also typicallyincludes a single receive chain (Rx) 228, 328 that includes I and Qanalog to digital converters (ADCs), one or more combinations of an LOand a frequency synthesizer, one or more channel filters, optionaldownconverters, a vector demodulator and an automatic gain control (AGC)amplifier. Note that in many cases some of the one or more LO andfrequency synthesizer combinations can be shared between the Tx and Rxchains.

As shown in FIGS. 2 and 3, conventional PTP radios 200, 300 also includea single power amplifier (PA) 232, 332. The PA 232, 332 boosts thetransmit signal to a level appropriate for radiation from the antenna inkeeping with relevant regulatory restrictions and instantaneous linkconditions. Similarly, such conventional PTP radios 232, 332 typicallyalso include a single low-noise amplifier (LNA) 236, 336 as shown inFIGS. 2 and 3. The LNA 236, 336 boosts the received signal at theantenna while minimizing the effects of noise generated within theentire signal path.

As described above, FIG. 2 illustrates a conventional PTP radio 200 forthe case of TDD operation. As shown in FIG. 2, conventional PTP radios200 typically connect the antenna 240 to the PA 232 and LNA 236 via aband-select filter 244 and a single-pole, single-throw (SPST) switch242.

As described above, FIG. 3 illustrates a conventional PTP radio 300 forthe case of FDD operation. As shown in FIG. 3, in conventional PTPradios 300, then antenna 340 is typically connected to the PA 332 andLNA 336 via a duplexer filter 344. The duplexer filter 344 isessentially two band-select filters (tuned respectively to the Tx and Rxbands) connected at a common point.

In the conventional PTP radios shown in FIGS. 2 and 3, the antenna 240,340 is typically of very high gain such as can be achieved by aparabolic dish so that gains of typically >30 dBi (or even sometimes >40dBi), can be realized. Such an antenna usually has a narrow radiationpattern in both the elevation and azimuth directions. The use of such ahighly directive antenna in a conventional PTP radio link withunobstructed LOS propagation conditions ensures that the modem 216, 316has insignificant impairments at the receiver (antenna 240, 340) due tomultipath self-interference and further substantially reduces thelikelihood of unwanted co-channel interference due to other nearby radiolinks.

Although not explicitly shown in FIGS. 2 and 3, the conventional PTPradio may use a single antenna structure with dual antenna feedsarranged such that the two electromagnetic radiation patterns emanatedby such an antenna are nominally orthogonal to each other. An example ofthis arrangement is a parabolic dish. Such an arrangement is usuallycalled dual-polarized and can be achieved either by orthogonal verticaland horizontal polarizations or orthogonal left-hand circular andright-hand circular polarizations.

When duplicate modem blocks, RF blocks, and PA/LNA/switch blocks areprovided in a conventional PTP radio, then connecting each PHY chain toa respective polarization feed of the antenna allows theoretically up totwice the total amount of information to be communicated within a givenchannel bandwidth to the extent that cross-polarizationself-interference can be minimized or cancelled sufficiently. Such asystem is said to employ “dual-polarization” signaling.

When an additional circuit (not shown) is added to FIG. 2 that canprovide either the RF Tx signal or its anti-phase equivalent to eitherone or both of the two polarization feeds of such an antenna, then“cross-polarization” signaling can be used to effectively expand theconstellation of the modem within any given symbol rate or channelbandwidth. With two polarizations and the choice of RF signal or itsanti-phase, then an additional two information bits per symbol can becommunicated across the link. Theoretically, this can be extended andexpanded to additional phases, representing additional information bits.At the receiver, for example, a circuit (not shown) could detect if thetwo received polarizations are anti-phase with respect to each other, ornot, and then combine appropriately such that the demodulator in themodem block can determine the absolute phase and hence deduce the valuesof the two additional information bits. Cross-polarization signaling hasthe advantage over dual-polarization signaling in that it is generallyless sensitive to cross-polarization self-interference but for highorder constellations such as 64-QAM or 256-QAM, the relative increase inchannel efficiency is smaller.

In the conventional PTP radios shown in FIGS. 2 and 3, substantially allthe components are in use at all times when the radio link is operative.However, many of these components have programmable parameters that canbe controlled dynamically during link operation to optimize throughoutand reliability for a given set of potentially changing operatingconditions. The conventional PTP radios of FIGS. 2 and 3 control theselink parameters via a Radio Link Controller (RLC) 248, 348. The RLCfunctionality is also often described as a Link Adaptation Layer that istypically implemented as a software routine executed on amicrocontroller within the radio that can access the MAC 212, 312, Modem216, 316, RF 220, 320 and/or possibly other components with controllableparameters. The RLC 248, 348 typically can both vary parameters locallywithin its radio and communicate with a peer RLC at the other end of theconventional PTP radio link via “control frames” sent by the MAC 212,312 with an appropriate identifying field within a MAC Header 404 (inreference to FIG. 4).

Typical parameters controllable by the RLC 248, 348 for the Modem 216,316 of a conventional PTP radio include encoder type, encoding rate,constellation selection and reference symbol scheduling and proportionof any given PHY Protocol Data Unit (PPDU). Typical parameterscontrollable by the RLC 248, 348 for the RF 220, 320 of a conventionalPTP radio include channel frequency, channel bandwidth, and output powerlevel. To the extent that a conventional PTP radio employs twopolarization feeds within its single antenna, additional parameters mayalso be controlled by the RLC 248, 348 as self-evident from thedescription above.

In conventional PTP radios, the RLC 248, 348 decides, usuallyautonomously, to attempt such parameter changes for the link in responseto changing propagation environment characteristics such as, forexample, humidity, rain, snow, or co-channel interference. There areseveral well-known methods for determining that changes in thepropagation environment have occurred such as monitoring the receivesignal strength indicator (RSSI), the number of or relative rate of FCSfailures at the MAC 212, 312, and/or the relative value of certaindecoder accuracy metrics. When the RLC 248, 348 determines thatparameter changes should be attempted, it is necessary in most casesthat any changes at the transmitter end of the link become known to thereceiver end of the link in advance of any such changes. Forconventional PTP radios, and similarly for many other radios, there areat least two well-known techniques which in practice may not be mutuallyexclusive. First, the RLC 248, 348 may direct the PHY, usually in theModem 216, 316 relative to FIGS. 2 and 3, to pre-pend a PHY layerconvergence protocol (PLCP) header to a given PPDU that includes one ormore (or a fragment thereof) given MPDUs wherein such PLCP header hasinformation fields that notify the receiving end of the link ofparameters used at the transmitting end of the link. Second, the RLC248, 348 may direct the MAC 212, 312 to send a control frame, usually toa peer RLC 248, 348, including various information fields that denotethe link adaptation parameters either to be deployed or to be requestedor considered.

The foregoing describes at an overview level the typical structural andoperational features of conventional PTP radios which have been deployedin real-world conditions for many radio links where unobstructed (orsubstantially unobstructed) LOS propagation was possible. Theconventional PTP radio on a whole is completely unsuitable forobstructed LOS or PMP operation.

SUMMARY

The following summary of the invention is included in order to provide abasic understanding of some aspects and features of the invention. Thissummary is not an extensive overview of the invention and as such it isnot intended to particularly identify key or critical elements of theinvention or to delineate the scope of the invention. Its sole purposeis to present some concepts of the invention in a simplified form as aprelude to the more detailed description that is presented below.

According to an aspect of the invention, a fixed wireless access radiofor exchanging one or more data interface streams with one or more otherfixed wireless access radios is disclosed that includes a plurality ofreceive radio frequency (RF) chains, wherein at least a first subset ofthe plurality of receive RF chains is configured to convert from atleast a respective one of a plurality of receive RF signals within atleast a first receive frequency band to a respective one of a firstplurality of receive chain output signals, and wherein at least a secondsubset of the plurality of receive RF chains is configured to convertfrom at least a respective one of a plurality of receive RF signalswithin at least a second receive frequency band to a respective one of asecond plurality of receive chain output signals; a plurality oftransmit radio frequency (RF) chains, wherein at least a first subset ofthe plurality of transmit RF chains is configured to convert from atleast a respective one of a first plurality of transmit chain inputsignals to a respective one of a plurality of transmit RF signals withina first transmit frequency band, and wherein at least a second subset ofthe plurality of transmit RF chains is configured to convert from atleast a respective one of a second plurality of transmit chain inputsignals to a respective one of a plurality of transmit RF signals withina second transmit frequency band; and a plurality of antenna elements,wherein at least a first subset of the plurality of antenna elements isconfigured to operate over at least both of the first transmit frequencyband and the first receive frequency band and each antenna element ofthe first subset of the plurality of antenna elements is coupled orcouplable to at least one of the first subset of the plurality ofreceive RF chains or coupled or couplable to at least one of the firstsubset of the plurality of transmit RF chains, and wherein at least asecond subset of the plurality of antenna elements is configured tooperate over at least both of the second transmit frequency band and thesecond receive frequency band and each antenna element of the secondsubset of the plurality of antenna elements is coupled or couplable toat least one of the second subset of the plurality of receive RF chainsor coupled or couplable to at least one of the second subset of theplurality of transmit RF chains; wherein the radio is configured toprovide a base throughput capability using the first receive frequencyband and the first transmit frequency band; and wherein the radio isfurther configured to provide a surge throughput capability using thesecond receive frequency band and the second transmit frequency band.

The first transmit frequency band may be coincident with the firstreceive frequency band.

A first frequency band may include at least the first transmit frequencyband and the first receive frequency band.

The first frequency band may be either within a frequency range ofbetween 2 GHz and 7 GHz or within a frequency range of above 10 GHz.

The second transmit frequency band may be coincident with the secondreceive frequency band.

A second frequency band includes at least the second transmit frequencyband and the second receive frequency band.

The second frequency band may be either within a frequency range ofbetween 2 GHz and 7 GHz or within a frequency range of above 10 GHz.

At least one antenna element of the plurality of antenna elements may bea directive gain antenna element.

At least one antenna element of the plurality of antenna elements may beat least one of a patch antenna element, a dipole antenna element, or aslot antenna element.

At least one antenna element of the plurality of antenna elements may becoupled or couplable to at least one receive RF chain or transmit RFchain via either at least one RF switch or at least one duplexer filter.

A transmit path modulation format may be based upon Single-CarrierFrequency Domain Equalization (SC-FDE). A transmit path modulationformat may be based upon Orthogonal Frequency Division Multiplexing(OFDM).

The fixed wireless access radio may be configured to transmit in thefirst transmit frequency band and receive in the first receive frequencyband coincident in time for at least a period of time.

The fixed wireless access radio may be configured to transmit in thesecond transmit frequency band and receive in the second receivefrequency band coincident in time for at least a period of time.

The surge throughput capability may be higher than the base throughputcapability.

The risk of temporal interference outage for the surge throughputcapability may be higher than for the base throughput capability.

At least one of the plurality of receive RF chains includes at least avector demodulator and two analog to digital converters that areconfigured to produce a respective one of a plurality of receive chainoutput signals comprised of digital baseband quadrature signals.

At least one of the plurality of transmit RF chains includes at least avector modulator and two digital to analog converters that areconfigured to produce a respective one of the plurality of transmit RFsignals from a respective one of a plurality of transmit chain inputsignals comprised of digital baseband quadrature signals.

In accordance with another aspect of the invention, a fixed wirelessaccess radio for exchanging one or more data interface streams with oneor more other fixed wireless access radios is disclosed that includes aplurality of receive radio frequency (RF) chains, wherein each of theplurality of receive RF chains is configured to convert from arespective one of a plurality of receive RF signals within a receivefrequency band to a respective one of a plurality of receive chainoutput signals; a plurality of transmit radio frequency (RF) chains,wherein each of the plurality of transmit RF chains is configured toconvert from a respective one of a plurality of transmit chain inputsignals to a respective one of a plurality of transmit RF signals withina transmit frequency band; a plurality of directive gain antennaelements, wherein each of the plurality of directive gain antennaelements is configured to operate over at least both of the transmitfrequency band and the receive frequency band; and a plurality ofduplexer filters, wherein each duplexer filter comprises at least areceive band-select filter configured to selectively pass RF signalswithin the receive frequency band and a transmit band-select filterconfigured to selectively pass RF signals within the transmit frequencyband, wherein each duplexer filter is couplable or coupled to at leastone of the plurality of directive gain antenna elements, wherein thereceive band-select filter of each duplexer filter is couplable orcoupled to at least one of the plurality of receive RF chains, andwherein the transmit band-select filter of each duplexer filter iscouplable or coupled to at least one of the plurality of transmit RFchains; wherein the fixed wireless access radio is configured to operateat least a first subset of the plurality of transmit RF chains at afirst transmit RF carrier frequency and to operate at least a secondsubset of the plurality of transmit RF chains at a second transmit RFcarrier frequency; and wherein the fixed wireless access radio isfurther configured to select at least one of the first transmit RFcarrier frequency or the second transmit RF carrier frequency inresponse to at least a current link condition at an at least one of theone or more other fixed wireless access radios.

The fixed wireless access radio may further include one or moredemodulator cores, wherein each demodulator core is configured todemodulate one or more of a plurality of receive symbol streams toproduce one or more receive data interface streams; and a frequencyselective receive path channel multiplexer, interposed between the oneor more demodulator cores and at least the plurality of receive RFchains, wherein the frequency selective receive path channel multiplexeris configured to generate the plurality of receive symbol streams fromat least the plurality of receive chain output signals.

Each one of the one or more demodulator cores includes at least adecoder and a soft decision symbol demapper; and wherein each one of theplurality of receive RF chains comprises at least a vector demodulatorand two analog to digital converters that are configured to produce therespective one of the plurality of receive chain output signals, eachsaid respective one of the plurality of receive chain output signalscomprised of digital baseband quadrature signals.

Each one of the plurality of transmit RF chains includes at least avector modulator and two digital to analog converters that areconfigured to produce the respective one of the plurality of transmit RFsignals, each said respective one of the plurality of transmit chaininput signals comprised of digital baseband quadrature signals.

Each one of the one or more demodulator cores may include at least oneof a descrambler or a deinterleaver; and wherein each one of the one ormore modulator cores comprises at least one of a scrambler or aninterleaver.

The fixed wireless access radio may further include one or moreselectable RF connections that are configured to selectively couplecertain of the plurality of directive gain antenna elements to either orboth of certain of the plurality of receive RF chains or certain of theplurality of transmit RF chains; wherein the number of directive gainantenna elements that are configured to be selectively coupled toreceive RF chains exceeds the number of receive RF chains that areconfigured to accept receive RF signals from the one or more selectableRF connections; or wherein the number of directive gain antenna elementsthat are configured to be selectively coupled to transmit RF chainsexceeds the number of transmit RF chains that are configured to providetransmit RF signals to the one or more selectable RF connections.

At least one of the one or more selectable RF connections my include atleast one RF switch.

The set of receive RF chains that is configured to accept receive RFsignals from the one or more selectable RF connections may be dividedbetween a first subset that is configured to accept receive RF signalsfrom directive gain antenna elements with a first polarization and asecond subset that is configured to accept receive RF signals fromdirective gain antenna elements with a second polarization; or whereinthe set of transmit RF chains that is configured to provide transmit RFsignals to the one or more selectable RF connections is divided betweena third subset that is configured to provide transmit RF signals todirective gain antenna elements with a first polarization and a fourthsubset that is configured to provide transmit RF signals to directivegain antenna elements with a second polarization.

The directive gain antenna elements may be arranged on a plurality offacets with one or more directive gain antenna elements per facet, andwherein each facet is oriented at a different azimuth angle relative toat least one other facet.

The fixed wireless access radio may further include a plurality of poweramplifiers, wherein each power amplifier is configured to amplify atleast one of the transmit RF signals, and wherein each power amplifieris couplable or coupled to at least one of the plurality of transmit RFchains and to at least one transmit band-select filter of the pluralityof duplexer filters; and a plurality of low noise amplifiers, whereineach low noise amplifier is configured to amplify at least one of thereceive RF signals, and wherein each low noise amplifier is couplable orcoupled to at least one of the plurality of receive RF chains and to atleast one receive band-select filter of the plurality of duplexerfilters.

The first transmit frequency band may be coincident with the firstreceive frequency band.

The first frequency band may include at least the first transmitfrequency band and the first receive frequency band.

The first frequency band may be either within a frequency range ofbetween 2 GHz and 7 GHz or within a frequency range of above 10 GHz.

The second transmit frequency band may be coincident with the secondreceive frequency band.

The second frequency band may include at least the second transmitfrequency band and the second receive frequency band.

The second frequency band may be either within a frequency range ofbetween 2 GHz and 7 GHz or within a frequency range of above 10 GHz.

The frequency selective receive path channel multiplexer may include atleast one of a Space Division Multiple Access (SDMA) combiner orequalizer, a maximal ratio combining (MRC) combiner or equalizer, aminimum mean squared error (MMSE) combiner or equalizer, an Eigen BeamForming (EBF) combiner or equalizer, a receive beam forming (BF)combiner or equalizer, a Zero Forcing (ZF) combiner or equalizer, achannel estimator, a Maximal Likelihood (DL) detector, an InterferenceCanceller (IC), a VBLAST combiner or equalizer, a Discrete FourierTransformer (DFT), a Fast Fourier Transformer (FFT), or an Inverse FastFourier Transformer (IFFT).

The frequency selective receive path channel multiplexer may include aplurality of cyclic prefix removers, wherein each cyclic prefix removeris configured to discard a fraction of an overall number of sampleswithin one or more blocks of a plurality of blocks of samples from arespective one of the plurality of receive chain output signals toproduce a respective cyclic prefix removed one or more blocks ofsamples, said fraction corresponding to a known cyclic prefix length fora plurality of second transmit symbol streams expected to be comprisedwithin the plurality of receive chain output signals; a plurality ofrespective complex Discrete Fourier Transformers coupled to eachrespective cyclic prefix remover, wherein each complex Discrete FourierTransformer is configured to decompose the respective cyclic prefixremoved one or more blocks of samples into a respective set of receivechain frequency domain subchannel samples; and a plurality of receivechannel equalizers coupled to the plurality of respective complexDiscrete Fourier Transformers, wherein each receive channel equalizer isconfigured to produce a set of channel-equalized frequency domainestimates representative of a respective one of the plurality of secondtransmit symbol streams by applying respective stream-specific andchain-specific receive weights to the respective sets of receive chainfrequency domain subchannel samples; wherein said respectivestream-specific and chain-specific receive weights applied to therespective sets of receive chain frequency domain subchannel samplesvary with relative frequency domain subchannel position within suchsets.

The fixed wireless access radio may further include a channel equalizercoefficients generator, wherein the channel equalizer coefficientsgenerator is configured to determine the respective stream-specific andchain-specific receive weights based at least upon comparison of certainsets of receive chain frequency domain subchannel samples with certainexpected blocks of known frequency domain subchannel samples expected tobe present at certain times within the plurality of receive chain outputsignals.

The fixed wireless access radio may further include a plurality ofcomplex Inverse Discrete Fourier Transformers, wherein each complexInverse Discrete Fourier Transformer is configured to compose arespective one of the plurality of receive symbol streams fromrespective sets of channel-equalized frequency domain estimatesrepresentative of the respective one of the plurality of second transmitsymbol streams.

Each of the plurality of complex Inverse Discrete Fourier Transformersmay be implemented by a structure executing a complex Inverse FastFourier Transform (IFFT), and wherein each of the plurality of complexDiscrete Fourier Transformers is implemented by a structure executing acomplex Fast Fourier Transform (FFT).

Each of the plurality of receive channel equalizers may include a numberof complex multipliers corresponding to a number of the plurality ofreceive chain output signals, and a combiner.

A transmit path modulation format may be based upon Single-CarrierFrequency Domain Equalization (SC-FDE). A transmit path modulationformat may be based upon Orthogonal Frequency Division Multiplexing(OFDM).

The fixed wireless access radio may further include one or moremodulator cores, wherein each modulator core is configured to modulateone or more transmit data interface streams to produce one or more of aplurality of transmit symbol streams, wherein each transmit symbolstream comprises at least a plurality of blocks of symbols, and whereineach one of the one or more modulator cores comprises at least anencoder and a symbol mapper; a non-frequency selective transmit pathchannel multiplexer, interposed between the one or more modulator coresand at least the plurality of transmit RF chains, wherein thenon-frequency selective transmit path channel multiplexer is configuredto generate the plurality of transmit chain input signals from at leastthe plurality of transmit symbol streams; wherein a first number of theplurality of transmit chain input signals exceeds a second number of theplurality of transmit symbol streams; and wherein the non-frequencyselective transmit path channel multiplexer is configured to applyrespective sets of stream-specific and chain-specific transmitbeamforming weights to at least one or more blocks of the plurality ofblocks of symbols from the plurality of transmit symbol streams whengenerating a respective one of the plurality of transmit chain inputsignals, and wherein a particular one of said stream-specific andchain-specific transmit beamforming weights is invariant with respect toa relative symbol position within said at least one or more blocks ofthe plurality of blocks of symbols.

The non-frequency selective transmit path channel multiplexer mayinclude a plurality of cyclic prefix adders, wherein each cyclic prefixadder is configured to add a fraction of an overall number of sampleswithin one or more blocks of a plurality of blocks of samplescorresponding to a respective one of the plurality of transmit chaininput signals, said fraction corresponding to a pre-determined cyclicprefix length; and a plurality of transmit channel equalizers, whereineach transmit channel equalizer is configured to produce one or moreblocks of non-frequency selective, channel-equalized samplescorresponding to a respective one of the plurality of transmit chaininput signals by applying respective sets of the stream-specific andchain-specific transmit beamforming weights to corresponding blocks ofsymbols from the plurality of transmit symbol streams; wherein a numberof the plurality of cyclic prefix adders and of the plurality oftransmit channel equalizers corresponds to the first number.

The fixed wireless access radio may further include a plurality ofcomplex Inverse Discrete Fourier Transformers, wherein each complexInverse Discrete Fourier Transformer is configured to compose arespective one of the plurality of transmit chain input signals fromrespective ones of non-frequency selective, channel-equalized samplescorresponding to respective ones of the plurality of transmit chaininput signals.

An output from each respective one of the plurality of transmit channelequalizers may be coupled to an input of a respective one of theplurality of cyclic prefix adders.

Each of the plurality of transmit channel equalizers may include anumber of complex multipliers corresponding to the second number, and acombiner.

The stream-specific and chain-specific transmit beamforming weights aredetermined at a receiver comprised within at least one of the fixedwireless access radio or the one or more other fixed wireless accessradios.

The receiver that determines the stream-specific and chain-specifictransmit beamforming weights may further include a channel equalizercoefficients generator, wherein the channel equalizer coefficientsgenerator is configured to determine the respective stream-specific andchain-specific transmit beamforming weights based at least uponcomparison of certain signals at the receiver with certain expectedsignals expected to be present at certain times.

The stream-specific and chain-specific transmit beamforming weights maybe determined in order to improve either a signal to interference andnoise ratio (SINR) or a signal to noise ratio (SNR).

Each of the stream-specific and chain-specific transmit beamformingweights includes at least a real branch component and an imaginarybranch component.

Each of the stream-specific and chain-specific transmit beamformingweights includes at least one of an amplitude component or a phasecomponent.

A transmit path modulation format may be based upon Single-CarrierFrequency Domain Equalization (SC-FDE). A transmit path modulationformat may be based upon Orthogonal Frequency Division Multiplexing(OFDM).

The fixed wireless access radio may further include a radio resourcecontroller (RRC); wherein the radio resource controller is configured toselect the at least one of the first transmit RF carrier frequency orthe second transmit RF carrier frequency in response to at least thecurrent link condition at the at least one of the one or more otherfixed wireless access radios.

A first channel bandwidth corresponding to the first transmit RF carrierfrequency may be equal to a second channel bandwidth corresponding tothe second transmit RF carrier frequency.

A first channel bandwidth corresponding to the first transmit RF carrierfrequency may be not equal to a second channel bandwidth correspondingto the second transmit RF carrier frequency.

The current link condition may be derived from at least one or more linkquality metrics determined at the at least one of the one or more otherfixed wireless access radios.

At least one or more link quality metrics may include at least one ormore of a receive strength signal indication (RSSI), a decoder metric, aframe check sum (FCS) failure rate, a signal to noise ratio (SNR) or asignal to interference and noise ratio (SINR).

The fixed wireless access radio may be configured to operate at least afirst subset of the plurality of receive RF chains at a first receive RFcarrier frequency and to operate at least a second subset of theplurality of receive RF chains at a second receive RF carrier frequency.

In accordance with a further aspect of the invention, a fixed wirelessaccess radio for exchanging one or more data interface streams with oneor more other fixed wireless access radios is disclosed that includes aplurality of receive radio frequency (RF) chains, wherein each of theplurality of receive RF chains is configured to convert from arespective one of a plurality of receive RF signals within a receivefrequency band to a respective one of a plurality of receive chainoutput signals; a plurality of transmit radio frequency (RF) chains,wherein each of the plurality of transmit RF chains is configured toconvert from a respective one of a plurality of transmit chain inputsignals to a respective one of a plurality of transmit RF signals withina transmit frequency band; and a plurality of directive gain antennaelements, wherein a first subset of the plurality of directive gainantenna elements is configured to operate over at least the transmitfrequency band and is couplable or coupled to at least one the pluralityof transmit RF chains and a second subset of the plurality of directivegain antenna elements is configured to operate over at least the receivefrequency band and is couplable or coupled to at least one the pluralityof receive RF chains; wherein the fixed wireless access radio isconfigured to operate at least a first subset of the plurality oftransmit RF chains at a first transmit RF carrier frequency and tooperate at least a second subset of the plurality of transmit RF chainsat a second transmit RF carrier frequency; and wherein the fixedwireless access radio is further configured to select at least one ofthe first transmit RF carrier frequency or the second transmit RFcarrier frequency in response to at least a current link condition at anat least one of the one or more other fixed wireless access radios.

Each of the plurality of directive gain antenna elements may beconfigured to operate over at least both of the transmit frequency bandand the receive frequency band.

The transmit frequency band may be coincident with the receive frequencyband.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are incorporated into and constitute apart of this specification, illustrate one or more examples ofembodiments and, together with the description of example embodiments,serve to explain the principles and implementations of the embodiments.

FIG. 1 is an illustration of conventional point to point (PTP) radiosdeployed for cellular base station backhaul with unobstructed line ofsight (LOS).

FIG. 2 is a block diagram of a conventional PTP radio for Time DivisionDuplex (TDD).

FIG. 3 is a block diagram of a conventional PTP radio for FrequencyDivision Duplex (FDD).

FIG. 4 is an illustration of a MAC Protocol Data Unit (MPDU).

FIG. 5 is an illustration of intelligent backhaul radios (IBRs) deployedfor cellular base station backhaul with obstructed LOS according to oneembodiment of the invention.

FIG. 6 is a block diagram of an IBR according to one embodiment of theinvention.

FIG. 7 is a block diagram of an IBR according to one embodiment of theinvention.

FIG. 8 is a block diagram illustrating an exemplary deployment of IBRsaccording to one embodiment of the invention.

FIG. 9 is a block diagram illustrating an exemplary deployment of IBRsaccording to one embodiment of the invention.

FIG. 10 is a block diagram of an IBR antenna array according to oneembodiment of the invention.

FIG. 11 is a block diagram of a front-end unit for TDD operationaccording to one embodiment of the invention.

FIG. 12 is a block diagram of a front-end unit for FDD operationaccording to one embodiment of the invention.

FIG. 13 is a perspective view of an IBR according to one embodiment ofthe invention.

FIG. 14 is a perspective view of an IBR according to one embodiment ofthe invention.

FIG. 15 is a perspective view of an IBR according to one embodiment ofthe invention.

FIG. 16 is a block diagram illustrating an exemplary transmit chainwithin an IBR RF according to one embodiment of the invention.

FIG. 17 is a block diagram illustrating an exemplary receive chainwithin an IBR RF according to one embodiment of the invention.

FIG. 18 is a block diagram illustrating an IBR modem according to oneembodiment of the invention.

FIG. 19 is a block diagram illustrating a modulator core j according toone embodiment of the invention.

FIG. 20 is a block diagram illustrating a demodulator core j accordingto one embodiment of the invention.

FIG. 21 is a block diagram illustrating a modulator core j according toone embodiment of the invention.

FIG. 22 is a block diagram illustrating a demodulator core j accordingto one embodiment of the invention.

FIG. 23, consisting of FIG. 23A and FIG. 23B, is a block diagramillustrating a channel multiplexer (MUX) according to one embodiment ofthe invention. FIG. 23A is a partial view showing the transmit path andthe channel equalizer coefficients generator within the exemplarychannel MUX. FIG. 23B is a partial view showing the receive path withinthe exemplary channel MUX.

FIG. 24 is a block diagram illustrating an exemplary Tx-CE-m accordingto one embodiment of the invention.

FIG. 25 is a block diagram illustrating an exemplary Rx-CE-l accordingto one embodiment of the invention.

FIG. 26 is a timing diagram illustrating processing of PPDU-l withTx-path and Rx-path of respective IBR channel MUXs according to oneembodiment of the invention.

FIG. 27 is a block diagram illustrating an exemplary Tx PLCP accordingto one embodiment of the invention.

FIG. 28 is a block diagram illustrating an exemplary Tx PLCP Mod-jaccording to one embodiment of the invention.

FIG. 29 is a block diagram illustrating an exemplary Rx PLCP accordingto one embodiment of the invention.

FIG. 30 is a block diagram illustrating an exemplary Rx PLCP Mod-jaccording to one embodiment of the invention.

FIG. 31 is a schematic diagram of an IBR communications protocols stackaccording to one embodiment of the invention.

FIG. 32 is a schematic diagram of an IBR communications protocols stackaccording to one embodiment of the invention.

FIG. 33 is a block diagram of an IBR media access control (MAC)according to one embodiment of the invention.

FIG. 34 is a timing diagram illustrating channel activity for FDD withfixed superframe timing according to one embodiment of the invention.

FIG. 35 is a timing diagram illustrating channel activity for TDD withfixed superframe timing according to one embodiment of the invention.

FIG. 36 is a timing diagram illustrating channel activity for TDD/CSMAwith variable superframe timing according to one embodiment of theinvention.

DETAILED DESCRIPTION

FIG. 5 illustrates deployment of intelligent backhaul radios (IBRs) inaccordance with an embodiment of the invention. As shown in FIG. 5, theIBRs 500 are deployable at street level with obstructions such as trees504, hills 508, buildings 512, etc. between them. The IBRs 500 are alsodeployable in configurations that include point to multipoint (PMP), asshown in FIG. 5, as well as point to point (PTP). In other words, eachIBR 500 may communicate with more than one other IBR 500.

For 3G and especially for 4^(th) Generation (4G), cellular networkinfrastructure is more commonly deployed using “microcells” or“picocells.” In this cellular network infrastructure, compact basestations (eNodeBs) 516 are situated outdoors at street level. When sucheNodeBs 516 are unable to connect locally to optical fiber or a copperwireline of sufficient data bandwidth, then a wireless connection to afiber “point of presence” (POP) requires obstructed LOS capabilities, asdescribed herein.

For example, as shown in FIG. 5, the IBRs 500 include an Aggregation EndIBR (AE-IBR) and Remote End IBRs (RE-IBRs). The eNodeB 516 of the AE-IBRis typically connected locally to the core network via a fiber POP 520.The RE-IBRs and their associated eNodeBs 516 are typically not connectedto the core network via a wireline connection; instead, the RE-IBRs arewirelessly connected to the core network via the AE-IBR. As shown inFIG. 5, the wireless connection between the IBRs include obstructions(i.e., there may be an obstructed LOS connection between the RE-IBRs andthe AE-IBR). In an alternative embodiment, the IBRs shown in FIG. 5 arefixed wireless access radios or fixed wireless broadband access radios.

FIGS. 6 and 7 illustrate exemplary embodiments of the IBRs 500 shown inFIG. 5. In FIGS. 6 and 7, the IBRs 500 include interfaces 604, interfacebridge 608, MAC 612, modem 624, channel MUX 628, RF 632, which includesTx1 . . . TxM 636 and Rx1 . . . RxN 640, antenna array 648 (includesmultiple antennas 652), a Radio Link Controller (RLC) 656 and a RadioResource Controller (RRC) 660. The IBR may optionally include an IBMSagent 700 as shown in FIG. 7. It will be appreciated that the componentsand elements of the IBRs may vary from that illustrated in FIGS. 6 and7. The various elements of the IBR are described below by theirstructural and operational features in numerous different embodiments.

The external interfaces of the IBR (i.e., the IBR Interface Bridge 608on the wireline side and the IBR Antenna Array 648 (including antennas652) on the wireless side) are a starting point for describing somefundamental differences between the numerous different embodiments ofthe IBR 500 and the conventional PTP radios described above (or othercommonly known radio systems, such as those built to existing standardsincluding 802.11n (WiFi) and 802.16e (WiMax)).

In some embodiments, the IBR Interface Bridge 608 physically interfacesto standards-based wired data networking interfaces 604 as Ethernet 1through Ethernet P. “P” represents a number of separate Ethernetinterfaces over twisted-pair, coax or optical fiber. The IBR InterfaceBridge 608 can multiplex and buffer the P Ethernet interfaces 604 withthe IBR MAC 612. For the case of P=1 (a single Ethernet interface), themultiplexing operation of the IBR Interface Bridge 608 is a trivial“pass-through” between the single Ethernet interface and the buffer. Inexemplary embodiments, the IBR Interface Bridge 608 preserves “Qualityof Service” (QoS) or “Class of Service” (CoS) prioritization asindicated, for example, in IEEE 802.1q 3-bit Priority Code Point (PCP)fields within the Ethernet frame headers, such that either the IBR MAC612 schedules such frames for transmission according to policiesconfigured within the IBR of FIG. 6 or communicated via the IBMS Agent700 of FIG. 7, or the IBR interface bridge 608 schedules the transfer ofsuch frames to the IBR MAC 612 such that the same net effect occurs. Inother embodiments, the IBR interface bridge 608 also forwards andprioritizes the delivery of frames to or from another IBR over aninstant radio link based on Multiprotocol Label Switching (MPLS) orMultiprotocol Label Switching Transport Profile (MPLS-TP).

In some embodiments, the IBR Interface Bridge 608 can also perform layer2 switching of certain Ethernet interfaces to other Ethernet interfaces604 in response to radio link failure conditions and policies configuredwithin the IBR of FIG. 6 or communicated via the IBMS Agent 700 of FIG.7. A radio link failure condition can arise from any one or more ofmultiple possible events, such as the following exemplary radio linkfailure condition events:

physical failure of a component within the IBR other than the IBRInterface Bridge and its power supply;

degradation of the RF link beyond some pre-determined throughput leveldue to either changing propagation environment or additional co-channelinterference; and

failure of any kind at the other end of the RF link that preventsconnection to the ultimate source or destination.

FIG. 8 illustrates an exemplary configuration of multiple IBRs (IBR 1 1804, IBR 2 808). Each IBR 804, 808 has layer 2 switching in the IBRinterface bridges 816, 820 (each corresponding to an instance of the IBRInterface Bridge 608 of FIGS. 6 and 7). In one embodiment, the datanetworking traffic can be from, for example, a cellular eNodeB, a WiFihotspot, an enterprise edge router, or any of numerous other remote datanetworks 828. As shown in FIG. 8, the remote data network 828 isconnected via an Ethernet cable (copper or fiber) 832 to the firstEthernet port 836 on the IBR Interface Bridge 816 of IBR1 804. AnotherEthernet cable 840 connects the second Ethernet port 844 on the IBRInterface Bridge 816 of IBR1 804 to the first Ethernet port 848 on theIBR Interface Bridge 820 of IBR2 808. If a radio link failure conditionoccurs for any reason, such as those listed above, with respect to RFLink 1 852, then the layer 2 switch within the IBR Interface Bridge 816of IBR1 804 can automatically connect all data networking trafficoriginating from or destined to Remote Data Network 1 828 via IBR2 808and RF Link2 856, completely transparently to Remote Data Network 1 828.This provides fail over redundancy to reduce the probability of networkoutage at Remote Data Network 1 828.

In some embodiments, the IBR Interface Bridge with layer 2 switching canalso be configured to perform load balancing in response to operatingconditions and policies configured within the IBR of FIG. 6 orcommunicated via the IBMS Agent 700 of FIG. 7. For example, withreference to FIG. 8, the layer 2 switch in the IBR Interface Bridge 816of IBR1 804 can connect all data networking traffic in excess of datarates above a certain limit, such as a pre-determined level or theinstantaneous supportable rate on RF Link 1 852, over to IBR2 808. Forfull two-way functionality of this load balancing, an analogous loadbalancing capability exists within the layer 2 switch of an IBR at therespective other ends of both RF Link 1 852 and RF Link 2 856.

FIG. 9 illustrates an alternative configuration of IBRs 804, 808 withlayer 2 switching capability and optional load balancing capability forthe case of two disparate Remote Data Networks 1 and 2 (900 and 904,respectively). The two Remote Data Networks 1 and 2 (900 and 904) are ator within Ethernet cabling distance of two IBRs 1 and 2 (804, 808)operating on two RF links 1 and 2 (852, 856). As described above, theRemote Data Network 828 is connected via Ethernet cable 832 to the firstEthernet port 836, and the IBRs 1 and 2 (804, 808) are connected viaEthernet cable 840 at Ethernet ports 844 and 848 respectively. TheRemote Data Network 2 904 is connected to IBR 2 808 via Ethernet cable908 at the Ethernet port 912. In the embodiment shown in FIG. 9, if aradio link failure condition occurs for any reason, such as those listedabove, with respect to RF Link 1 852, then the IBR Interface Bridge 816of IBR1 804 can use its layer 2 switch to connect to the IBR InterfaceBridge 820 of IBR2 808 via Ethernet cable 840 such that IBR2 808 canbackhaul, subject to its throughput capabilities and prioritizationpolicies, the traffic originating from or destined to both Remote DataNetwork 1 900 and Remote Data Network 2 904. Similarly, the IBRs 804,808 can perform load balancing across both RF Links 1 and 2 (852, 856),for traffic originating from or destined to Remote Data Networks 1 and 2(900, 904).

In some embodiments, RF link 1 852 may utilize spectrum possessingadvantageous conditions, such as reduced interference, wider channelbandwidth, better propagation characteristics, and/or higher allowablepower than the spectrum utilized by RF Link 2 856, or vice versa. In thesituation where a radio link failure condition occurs with respect tothe more advantageous spectrum, either control signaling between the twoIBR Interface Bridges 816, 820 of IBRs 1 and 2 as shown in FIG. 6 ormessaging between the two IBMS Agents 700 as shown in FIG. 7, whetherdirectly or indirectly via one or more intermediaries, can cause theredundant IBR 808 to change to the advantageous spectrum no longer beingused by RF Link 852 with the failure condition.

FIG. 10 illustrates an exemplary embodiment of an IBR Antenna Array 648.FIG. 10 illustrates an antenna array having Q directive gain antennas652 (i.e., where the number of antennas is greater than 1). In FIG. 10,the IBR Antenna Array 648 includes an IBR RF Switch Fabric 1012, RFinterconnections 1004, a set of Front-ends 1008 and the directive gainantennas 652. The RF interconnections 1004 can be, for example, circuitboard traces and/or coaxial cables. The RF interconnections 1004 connectthe IBR RF Switch Fabric 1012 and the set of Front-ends 1008. EachFront-end 1008 is associated with an individual directive gain antenna652, numbered consecutively from 1 to Q.

FIG. 11 illustrates an exemplary embodiment of the Front-end circuit1008 of the IBR Antenna Array 648 of FIG. 10 for the case of TDDoperation, and FIG. 12 illustrates an exemplary embodiment of theFront-end circuit 1008 of the IBR Antenna Array 648 of FIG. 10 for thecase of FDD operation. The Front-end circuit 1008 of FIG. 11 includes atransmit power amplifier PA 1104, a receive low noise amplifier LNA1108, SPDT switch 1112 and band-select filter 1116. The Front-endcircuit 1008 of FIG. 12 includes a transmit power amplifier PA 1204,receive low noise amplifier LNA 1208, and duplexer filter 1212. Thesecomponents of the Front-end circuit are substantially conventionalcomponents available in different form factors and performancecapabilities from multiple commercial vendors.

As shown in FIGS. 11 and 12, each Front-end 1008 also includes an“Enable” input 1120, 1220 that causes substantially all active circuitryto power-down. Power-down techniques are well known. Power-down isadvantageous for IBRs in which not all of the antennas are utilized atall times. It will be appreciated that alternative embodiments of theIBR Antenna Array may not utilize the “Enable” input 1120, 1220 orpower-down feature. Furthermore, for embodiments with antenna arrayswhere some antenna elements are used only for transmit or only forreceive, then certain Front-ends (not shown) may include only thetransmit or only the receive paths of FIGS. 11 and 12 as appropriate.

As described above, each Front-end (FE-q) corresponds to a particulardirective gain antenna 652. Each antenna 652 has a directivity gain Gq.For IBRs intended for fixed location street-level deployment withobstructed LOS between IBRs, whether in PTP or PMP configurations, eachdirective gain antenna 652 may use only moderate directivity compared toantennas in conventional PTP systems at a comparable RF transmissionfrequency. Based on measurements of path loss taken at street level at2480 MHz in various locations in and nearby San Jose, Calif. duringAugust and September 2010, IBR antennas should have a Gq of at least 6dBi, and, in typical embodiments for operation between 2 GHz and 6 GHzRF transmission frequency, a Gq in the range of 10-18 dBi, wherein theradiation pattern in the elevation direction is typically less than 90°and nominally parallel to the local surface grade. It will beappreciated that the RF transmission frequency range may be greater than2 GHz and 6 GHz; for example, the RF transmission frequency may be inthe range of 2 GHz and 7 GHz. At higher RF transmission frequencies,higher gain ranges of Gq are expected to be preferable. For example, Gqmay be preferably 16-24 dBi for 20-40 GHz operation or 20-28 dBi for60-90 GHz operation. In one particular embodiment, the directive gainantennas 652 are “patch” type antennas with Gq of about 13 dBi andnominally equal elevation and azimuth radiation patterns (about 40°each). Patch type antennas are advantageous because they can be realizedusing substantially conventional printed circuit board (PCB) fabricationtechniques, which lowers cost and facilitates integration with Front-endcircuit components and/or some or substantially all of the IBR RF SwitchFabric. However, may other antenna types, such as helical, horn, andslot, as well as microstrip antennas other than patch (such as reflecteddipoles), and the like, may be used with the IBR Antenna Array. In analternative embodiment, the directive gain antennas 652 are reflecteddipoles with Gq of about 15 dBi (about 50° azimuth and 20° elevation).In many embodiments, the antenna elements are chosen with elevationangular patterns considerably less than azimuthal angular patterns.

In the IBR Antenna Array 648 illustrated in FIGS. 6, 7 and 10, the totalnumber of individual antenna elements 652, Q, is greater than or equalto the larger of the number of RF transmit chains 636, M, and the numberof RF receive chains 640, N. In some embodiments, some or all of theantennas 652 may be split into pairs of polarization diverse antennaelements realized by either two separate feeds to a nominally singleradiating element or by a pair of separate orthogonally orientedradiating elements. Such cross polarization antenna pairs enable eitherincreased channel efficiency or enhanced signal diversity as describedfor the conventional PTP radio. The cross-polarization antenna pairs aswell as any non-polarized antennas are also spatially diverse withrespect to each other.

In some embodiments, certain antenna elements 652 may be configured withdifferent antenna gain Gq and/or radiation patterns compared to othersin the same IBR to provide pattern diversity.

In some embodiments, some antenna elements 652 may be oriented indifferent ways relative to others to achieve directional diversity. Forexample, FIG. 13 illustrates an IBR suitable for obstructed LOS PTPoperation (or sector-limited PMP operation) in which spatial diversity(and optionally polarization diversity and/or pattern diversity) isutilized to the exclusion of directional diversity. As shown in FIG. 13,all of the antenna elements 1304 are positioned on a front facet 1308 ofthe IBR. In FIG. 13, the IBR 1300 includes eight antenna elements 1304(Q=8). It will be appreciated that the IBR 1300 may include less than ormore than eight antenna elements 1304.

FIG. 14 illustrates another embodiment of an IBR 1400 where directionaldiversity is present. IBR 1400 includes the same number of antennaelements as the IBR 1300 shown in FIG. 13 (Q=8, or 16 if usingcross-polarization feeds to all antenna elements). In FIG. 14, theantenna elements 1404 are arranged on a front facet 1408 and two sidefacets 1412. In FIG. 14, the side facets 1412 are at a 45° angle in theazimuth relative to the front facet 1408. It will be appreciated thatthis 45° angle is arbitrary and different angles are possible dependingon the specific radiation patterns of the various antenna elements.Furthermore, the angle may be adjustable so that the side facets 1412can vary in azimuth angle relative to the front facet between 0° to 90°(any value or range of values between 0° to 90°). Conventionalelectromechanical fabrication elements could also be used to make thisside facing angle dynamically adjustable by, for example, the RRC 660 ofFIG. 6 or the same in combination with the IBMS Agent 700 of FIG. 7.Additionally, variations of the embodiment of FIG. 14 can use more thanthree facets at different angular spacing all within a nominal azimuthalrange of approximately 180°, and the number of antenna elements 1404 maybe less than or greater than Q=8. For example, in one embodiment, theantenna array includes four facets uniformly distributed in an azimuthalangular range across 160°.

FIG. 15 illustrates an IBR 1500 having an “omni-directional” (in theazimuth) array of antenna elements 1504. In FIG. 15, Q=16 antennaelements 1504 are uniformly distributed across all 360° of azimuthangles, amongst the eight facets 1508-1536. Such an embodiment can beadvantageous for propagation environments with severe obstructionsbetween IBRs in a radio link or for an omni-directional common node at apoint of aggregation (i.e. fiber POP) within a PMP deployment of IBRs.It will be appreciated that the IBR may have less than or more thaneight facets, and that the number of antenna elements 1504 may be lessthan or greater than Q=16. It will also be appreciated that the antennaelements 1504 may be distributed non-uniformly across the facets.

With reference back to FIGS. 10-12, the IBR RF Switch Fabric 1012provides selectable RF connections between certain RF-Tx-m and/orRF-SW-Tx-q combinations and certain RF-Rx-n and/or RF-SW-Rx-qcombinations. In an embodiment where Q=M=N, the IBR RF Switch Fabric1012 can be parallel through connections or single-pole, single-throw(SPST) switches. In a maximally flexible embodiment where Q>Max (M, N)and any RF-Tx-m or RF-Rx-n can connect to any respective RF-SW-Tx-q orRF-SW-Rx-q, then a relatively complex cascade of individual switchblocks and a more extensive decoder logic may be required. Because eachRF-Tx-m or RF-Rx-n can be readily interchangeable amongst theirrespective sets of signals at the digital baseband level, it isgenerally only necessary to connect any given RF-Tx-m or RF-Rx-n to asubset of the Front-ends 1008 roughly by the ratio respectively of Q/Mor Q/N on average.

For example, if the IBR has Q=8 antenna elements and M=N=4, thenQ/M=Q/N=2. Thus, any of the RF-Tx-m (m=1, 2, 3, 4) signals may beconnectable to a pair of RF-SW-Tx-q signals, via a selectable RFconnection including a SPDT switch (and similarly for RF-Rx-n toRF-SW-Rx-q). In this example, either RF-Tx-m and/or RF-Rx-n couldconnect via such a selectable RF connection to either one of thefront-facing antenna elements or one of the side-facing antenna elementssuch that each RF signal has directional as well as spatial diversityoptions while allowing any two adjacent elements in the azimuthdirection to both be selected. Similarly, for the IBR shown in FIG. 15,if the IBR has Q=16 (non-polarized) antenna elements and M=N=4, then anyRF-Tx-n or RF-Rx-n signal could be oriented in one of four directions at90° increments via a selectable RF connection including a single-pole,quadrature throw (SP4T) switch.

An alternative embodiment of the IBR RF Switch Fabric 1012 can alsooptionally connect, via a signal splitter, a particular RF signal(typically one of the RF-Tx-m signals) to multiple Front-ends 1008 andantenna elements 652 simultaneously. This may be advantageous in someIBR operational modes to provide an effectively broader transmitradiation pattern either in normal operation or specifically for certainchannel estimation or broadcast signaling purposes. In context of theSPDT switch implementation in the example above for the IBR of FIG. 14,this would entail, if used for RF-Tx-m, the addition of another SPDTswitch and three passive splitter/combiners as well as decoder logic foreach antenna element pair.

In all of the foregoing descriptions of the IBR RF Switch Fabric 1012,substantially conventional components and RF board design structures asare well known can be used to physically implement such selectable RFconnections. Alternatively, these selectable RF connections can also berealized by custom integrated circuits on commercially-availablesemiconductor technologies.

With reference back to FIGS. 6 and 7, the IBR RF 632 also includestransmit and receive chains 636, 640. Exemplary transmit and receivechains 636, 640 are shown in FIGS. 16 and 17 respectively. In oneembodiment, as shown in FIG. 16, the transmit chain 636 takes a transmitchain input signal such as digital baseband quadrature signals I_(Tm)and Q_(Tm) and then converts them to a transmit RF signal RF-Tx-m.Typically, each transmit chain Tx-m 636 includes at least two signalDACs, channel filters, a vector modulator, a programmable gainamplifier, an optional upconverter, and at least one synthesized LO.Similarly, as shown in FIG. 17, the receive chain 640 converts a receiveRF signal RF-Rx-n to a receive chain output signal such as digitalbaseband quadrature signals I_(Rn) and Q_(Rn). Typically, each receivechain Rx-n 640 includes an optional downconverter, a vector demodulator,an AGC amplifier, channel filters, at least two signal ADCs and at leastone synthesized LO. A common synthesized LO can often be shared betweenpairs of Tx-m and Rx-n chains, or even amongst a plurality of suchpairs, for TDD operation IBRs. Examples of commercially availablecomponents to implement the IBR RF chains include the AD935x family fromAnalog Devices, Inc. Numerous other substantially conventionalcomponents and RF board design structures are well known as alternativesfor realizing the Tx-m and/or Rx-n chains whether for TDD or FDDoperation of the IBR.

With reference back to FIGS. 6 and 7, the specific details of the IBRModem 624 and IBR Channel MUX 628 depend somewhat on the specificmodulation format(s) deployed by the IBR. In general, the IBR requires amodulation format suitable for a broadband channel subject tofrequency-selective fading and multipath self-interference due to thedesired PHY data rates and ranges in obstructed LOS propagationenvironments. Many known modulation formats for such broadband channelsare possible for the IBR. Two such modulation formats for the IBR are(1) Orthogonal Frequency Division Multiplexing (OFDM) and (2)Single-Carrier Frequency Domain Equalization (SC-FDE). Both modulationformats are well known, share common implementation elements, and havevarious advantages and disadvantages relative to each other.

As is well known, OFDM essentially converts the frequency-selectivefading broadband channel into a parallel collection of flat-fadingsubchannels wherein the frequency spacing between subchannels is chosento maintain orthogonality of their corresponding time domain waveforms.In OFDM, a block of information symbols is transmitted in parallel on ablock of discrete frequency subchannels, each conveying one informationsymbol which can be efficiently channel multiplexed into the time domainby using an Inverse Discrete Fourier Transform (IDFT). A cyclic prefixof length in time greater than the dominant time delays associated withmulti-path self-interference is then pre-pended to the IDFT output blockby serially transmitting first in time a fraction of the IDFT outputblock time domain samples that are transmitted last. This length in timeis also sometimes called a guard interval. The use of a cyclic prefixeffectively converts a linear convolution of the transmitted block ofsymbols to a circular convolution such that the effects of inter-symbolinterference (ISI) associated with multipath time delays can be largelyeliminated at the OFDM receiver. At the OFDM receiver, the cyclic prefixis discarded and each time domain input block of symbols isdemultiplexed back into frequency domain subchannels each conveying oneinformation symbol by using a Discrete Fourier Transform (DFT). Thetransmission of a known training block of symbols within each OFDM PPDUenables the OFDM receiver to correct for carrier frequency offsets anddetermine a complex weighting coefficient for each frequency subchannelthat can equalize the effects of frequency-selective relative gain andphase distortion within the propagation channel. Furthermore,transmission of known “pilot” sequences of symbols at certainpredetermined subchannels within the transmit block enables the OFDMreceiver to track such channel distortions and frequency offsets duringreception of information symbol blocks the PPDU as well as provide acoherent reference for demodulation. Note that for those subchannelssubjected to severe flat fading, as will occur inevitably in a broadbandobstructed LOS propagation channel, the information within suchsubchannels cannot be directly demodulated. Thus to avoid a significantirreducible bit-error rate (BER) that would be unacceptable for most IBRapplications, it is essential that either forward error correction (FEC)encoding be applied with a constraint length comparable to the number ofbits per OFDM block of information symbols or with a combination ofconstraint length and interleaving depth such that related FEC encodedbits or symbols span substantially all of the OFDM block of informationsymbols.

In an SC-FDE transmitter, every block of information symbols, eachmapped to the same single carrier frequency at a relatively high symbolrate, has a cyclic prefix prepended to it prior to transmission. Similarto OFDM, the cyclic prefix consists of a finite fraction of themodulated symbols with a length in time greater than the dominant timedelays associated with multipath self-interference wherein suchmodulated symbols are identical to those to be transmitted last in timefor each block. Analogously to OFDM, this cyclic prefix effectivelyconverts a linear convolution of the transmitted block of symbols to acircular convolution such that inter-block interference (IBI) due tomultipath can be largely eliminated at the SC-FDE receiver. The SC-FDEreceiver is similar to the OFDM receiver in that a cyclic prefix removerdiscards a cyclic prefix for each block of information symbols and theremaining sampled signals are decomposed into a set of frequencysubchannels collectively representing the IBI-free block of symbolsusing a DFT. Based on a set of complex weighting coefficients, one foreach frequency sub-channel, as usually determined from a known trainingblock of symbols within each SC-FDE PPDU, the broadband channel inducedrelative distortions of amplitude and phase for each frequencysub-channel are then corrected in the Frequency Domain Equalizer (FDE).In contrast to OFDM where FDE-corrected subchannel estimates can bedirectly demapped as individual information symbols, in SC-FDE theFDE-corrected frequency domain subchannel estimates are thenre-multiplexed into a channel equalized single-carrier stream ofinformation symbol estimates using an IDFT so that such informationsymbol estimates can be subsequently demodulated.

Embodiments of the IBR may use Quadrature Amplitude Modulation (QAM) tomap groups of information data bits to a symbol including an I (or“real”) component and a Q (or “imaginary”) component. These symbols(i.e., symbols that include I and Q components) are typically referredto as “complex” symbols. Such “complex” symbols may be multiplexed ordemultiplexed by an IDFT or DFT respectively implemented by structuresexecuting a complex Inverse Fast Fourier Transform (IFFT) or complexFast Fourier Transform (FFT). In IBR embodiments, references to IDFT orDFT herein assume that such transforms will typically be implemented bystructures executing an IFFT or FFT respectively. Note also that thecyclic prefix described above can also be implemented as a cyclicpostfix for either OFDM or SC-FDE with equivalent performance. ForSC-FDE, some re-ordering of samples at the receiver after removal duringthe guard interval may be required if a cyclic postfix is used. It willbe appreciated however that techniques other than QAM for modulationmapping may also be used.

With reference again to FIGS. 6 and 7, the specific details of the IBRModem 624 and IBR Channel MUX 628 also depend somewhat on the specificantenna array signal processing format(s) deployed by the IBR. Ingeneral, the IBR utilizes multiple antennas and transmit and/or receivechains which can be utilized advantageously by several well-knownbaseband signal processing techniques that exploit multipath broadbandchannel propagation. Such techniques include Multiple-Input,Multiple-Output (MIMO), MIMO Spatial Multiplexing (MIMO-SM), beamforming(BF), maximal ratio combining (MRC), and Space Division Multiple Access(SDMA).

In general, any configuration where a transmitter that has multipletransmit antennas (or “inputs” to the propagation channel) communicateswith a receiver that has multiple receive antennas (or “outputs” fromthe propagation channel) can technically be described as MIMO. However,typically the term MIMO is used in the context of spatially multiplexingmultiple encoded and modulated information streams from multipletransmit antennas into a multipath channel for receipt at multiplereceive antennas wherein inversion of channel transfer matrix, usuallydetermined by a known training block associated with each PPDU, enablesseparation and demodulation/decoding of each information stream—aprocess called MIMO-SM. Various embodiments of the IBR, as describedherein, advantageously use other types of antenna diversity such asdirectional or polarization diversity to realize MIMO-SM performanceeven in propagation environments where spatial separation alone may beinadequate for conventional MIMO-SM.

For a given encoded and modulated information stream, BF or MRC can beutilized at either the transmitter or the receiver (or both) to improveeither the signal to interference and noise ratio (SINR) or the signalto noise ratio (SNR). For example, BF or MRC optimally combine theinformation signal received from the multiple antennas or split theinformation stream signal as transmitted by the multiple antennas.Numerous algorithms for determining the optimal weighting criteriaamongst the multiple antennas, usually as a function of frequency withina frequency-selective multipath broadband channel, are well known.

SDMA allows an Aggregation End IBR (AE-IBR) in a PMP configuration (seeFIG. 5) to transmit to or receive from multiple Remote End IBRs(RE-IBRs) simultaneously using parallel encoded and modulatedinformation streams each associated with multiple antenna andtransmit/receive chain combinations wherein stream-specific BF maximizessignal quality for a given stream and IBR pair, while minimizinginterference to other streams and IBR pairs. To the extent that multipleRE-IBRs are separable in space, or another antenna array characteristicsuch as polarization or direction, then significant increases in overallspectrum efficiency of a PMP system are possible using SDMA. As with BF(and MRC), numerous algorithms for computing the optimal weightingcriteria, such as Eigen Beam Forming (EBF), amongst multiple antennasare well known.

In view of the foregoing exemplary modulation format and antenna arrayprocessing format alternatives for the IBR, exemplary embodiments of theIBR Modem 624 and IBR Channel MUX 628 are described with reference toFIGS. 18-30.

FIG. 18 shows an exemplary embodiment of the IBR Modem 624. The IBRmodem 624 includes a Tx PLCP generator 1804, an Rx PLCP analyzer 1808,multiple modulator cores 1812 and multiple demodulator cores 1816. FIG.18 is divided functionally into a transmit path and a receive path. Thetransmit path includes the TxPLCP 1804 and the modulator cores 1812, andthe receive path includes the Rx PLCP analyzer 1808 and the demodulatorcores 1816.

The transmit path of the IBR Modem 624 includes a total of Jmod“Modulator Cores” 1812, each denoted as Modulator Core j wherein j=1, 2,. . . , Jmod. The Tx PLCP generator 1804 provides transmit datainterface streams, Tx-j, to each Modulator Core j 1812 such that Ajtotal vector outputs of mapped transmit symbol streams (denoted by the “

” on such I, Q connections) are generated from each Modulator Core j1812. This results in a total number of transmit symbol streams,K=A1+A2+ . . . +AJmod, each in vector format (I_(TS), Q_(TS))_(k) fromk=1 to K that are connected to the transmit path of the IBR Channel MUX628.

Similarly, the receive path of the IBR Modem 624 includes a total ofJdem “Demodulator Cores” 1816 each denoted as Demodulator Core j 1816wherein j=1, 2, . . . , Jdem. The IBR Channel MUX receive path providesL=B1+B2+ . . . +BJdem vector format receive symbol streams (I_(RS),Q_(RS))_(l) for l=1 to L that are input as Bj vector streams per eachDemodulator Core j 1816 to produce the receive data interface streamRx-j.

In PTP IBR configurations where Jmod=Jdem, usually Aj=Bj. However, for aPTP IBR where probing capability is present, Aj=Bj only for j=1 to Jmodin cases where Jdem>Jmod. In PMP IBR configurations, it may happen thatAj≠Bj; even if Jmod=Jdem.

An exemplary embodiment of a Modulator Core j 1812 is illustrated inFIG. 19. As shown in FIG. 19, this exemplary modulator core 1812includes a scrambler 1908, encoder 1912, stream parser 1916, multipleoptional interleavers 1920, multiple symbol groupers 1924 and multiplesymbol mappers 1928.

Typically, the data from the Tx PLCP generator 1804, Tx-j 1904, isscrambled at the scrambler 1908 and then passed to the encoder 1912. Theencoder 1912 provides FEC encoding and in some types of encoders alsoeffectively interleaves the encoded data. Exemplary FEC encoder typesfor IBRs include convolutional, turbo and low density parity check(LDPC). The encoded data is passed to the Stream Parser 1916. The StreamParser 1916 demultiplexes the encoded data into Aj streams. Each encodeddata stream is then interleaved if necessary at the optionalinterleavers 1920 such that the greater of the FEC encoder constraintlength and the interleaving depth approximates the total number of bitsper transmitted block of symbols in either OFDM or SC-FDE. Exampleinterleaver types include block (or “row/column”) and convolutional.Such interleaved and/or encoded data in each stream is then grouped atsymbol groupers 1924 based on the number of encoded bits per symbol. Thegroups of bits pass to the symbol mapper 1928, which converts each groupof encoded bits into a vector representation (I, Q) within aconstellation and then provides an output as a transmit symbol stream1932. An exemplary technique for mapping encoded bits to a vector symbolis QAM optionally with Gray coding.

FIG. 20 illustrates an exemplary embodiment of a Demodulator Core j 1816compatible with the exemplary Modulatory Core j 1812 of FIG. 19. Asshown in FIG. 20, the demodulator core 1816 includes a descrambler 2008,decoder 2012, stream MUX 2016, multiple optional deinterleavers 2020,and multiple soft decision symbol demappers 2024.

The Demodulator Core j 1816 at the highest level can be described asperforming essentially the reverse operations of those performed in theModulator Core j 1812 of FIG. 19. A key difference though is that thevector representation (I, Q) receive symbol streams 2032 input to eachDemodulator Core j 1816 are only estimates which may be corrupted due tochannel or receiver impairments such as multipath self-interference,Gaussian noise, co-channel interference, carrier frequency offset (CFO),distortion through channel filters at Tx and/or Rx, non-linearity in theTx and/or Rx chains and Front-ends, or phase noise in either Tx and/orRx local oscillators. Thus, the Demodulator Core j 1816 may use the softdecision symbol demapper 2024, which estimates the likelihood that areceived symbol or bit has a particular value using, for example, aknown technique such as Log-Likelihood Ratio (LLR). For example, if eachdata bit out of the demapper 2024 had a soft-decision representation (or“metric”) of 8 bits, then a value of 0 would represent a data bit of 0,a value of 1-20 would indicate most likely a data bit of 0, a value of21-125 would indicate more likely a data bit of 0 than a data bit of 1,a value of 126-129 would indicate near uncertainty of the data bit aseither 0 or 1, a value of 130-234 would indicate more likely a data bitof 1, a value of 235-254 would indicate most likely a data bit of 1, anda value of 255 would represent a data bit of 1. These soft-decisionmetrics can then be deinterleaved if applicable at optionaldeinterleavers 2020 and stream multiplexed at the Stream MUX 2016, andthen supplied to the decoder (deinterleaver) 2012 as a sequence ofsoft-decision metrics estimating the originally encoded (and possiblyinterleaved) bit stream. Decoder types are matched to encoder types asis well known. Techniques such as iterative decoding within the IBRmodem or combination of IBR modem and IBR Channel MUX are also known andcan be used in some embodiments.

FIGS. 19 and 20 are examples of implementations of the modulator anddemodulator cores 1812, 1816, respectively, in the IBR Modem 624. Itwill be appreciated that the order of the Scrambler 1908 and Encoder1912 can be reversed (and hence the Decoder 2012 and Descrambler 2008could be reversed). Also, the Stream Parser 1916 can divide a transmitdata interface stream Tx-j sequentially into multiple encoder/scramblercombinations and hence the corresponding Stream Mux 2016 would combinemultiple Decoder outputs. FIGS. 21 and 22 illustrate exemplaryalternatives with the above recited differences and without showingoptional separate interleavers 1920 and deinterleavers 2020.

Although each element is illustrated separately in FIGS. 19-22, theelements are not required to be distinct physical structures. It isfeasible to time multiplex either the entire Modulator or DemodulatorCores 1816, 1820 or constituent elements such as the Encoder 1912 orDecoder 2012 across multiple streams. Furthermore, in TDD operation, itcan be advantageous to time multiplex constituent elements such as theEncoder 1912 and/or Decoder 2012 even during times where the oppositetransmit/receive path is over the air compared to the time multiplexedelement. For example, by buffering receive symbol stream samples fromthe receive path of the IBR Channel MUX 628 during the receive portionof TDD operation, some of the streams can be decoded in a common Decoder2012 even when the IBR has changed over to transmit operation.

Note further that the exemplary embodiments of FIGS. 19-22 for theModulator and Demodulator Cores of the IBR Modem are all suitable foreither OFDM or SC-FDE operation.

FIG. 23 (FIGS. 23A and 23B together) illustrates an exemplary embodimentof the IBR Channel MUX 628. FIG. 23 is divided functionally into atransmit path (FIG. 23A) and a receive path (FIG. 23B). In FIG. 23A, thetransmit path includes multiple block assemblers 2304, multiple transmitchannel equalizers (CEs) 2308, multiple transmit MUXs 2312, multiplecyclic prefix addition (CPA) and block serializers 2316, multipletransmit digital front ends (DFEs) 2320, an optional preamble sampleslibrary 2324, and a pilot (and optionally preamble) symbol library 2328.In FIG. 23B, the receive path includes an acquisition/synchronizationcalculator 2336, multiple receive digital front ends (DFEs) 2340,multiple cyclic prefix removal (CPR) and block assemblers 2344, multiplecomplex DFTs 2348, multiple receive channel equalizers 2352 and multiplereceive MUXs 2356. The IBR Channel MUX 628 also includes a channelequalizer coefficients generator 2332 (see FIG. 23A). The channelequalizer coefficients generator 2332 may be used by both transmit andreceive paths or only the receive path depending on the specificembodiment. In the receive path, the IBR Channel MUX 628 is frequencyselective to enable operation in obstructed LOS propagation environmentswith frequency selective fading within the channel bandwidth as well asoperation in predominantly unobstructed LOS.

In the transmit path, each transmit symbol stream of mapped symbols(I_(TS), Q_(TS))_(k) from k=1 to K 2360 is connected to a respective TxBlock Assembler k 2304 as shown in FIG. 23A. For OFDM modulation, eachTx block Assembler k 2304 is equivalent to a serial to parallel bufferthat also places “running” pilot symbols in pre-determined locationscorresponding to pilot subchannels. In the exemplary embodiment of FIG.23A, such pilot symbols are supplied to each Tx Block Assembler k 2304by a Pilot Symbol Library 2328. Alternatively, pilot symbols areselectively injected into the data stream at predetermined points with aModulator Core 1812 similar to FIG. 19 or 21 but with a “Pilot DataLibrary” (not shown) providing grouped data to a selectable buffer ormultiplexer (not shown) between the Symbol Grouper 1924 and SymbolMapper 1928 of either FIG. 19 or 21.

For SC-FDE modulation with no frequency selective channel equalization,the Tx Block Assembler k 2304 would, as an exemplary embodiment, be asimple serial to parallel buffer that places pilot symbols inpre-determined symbol sequence positions either from the Pilot SymbolLibrary 2328 as shown, or alternatively from a “Pilot Data Library” inthe Modulator Core as described above. If frequency selective channelequalization in Tx for SC-FDE is used, then in addition to the above,the Tx Block Assembler k 2304 would also include a Complex DFT thatfollows the serial to parallel buffer. In this DFT structure version ofSC-FDE (also known as DFT pre-coding SC-DFE), it is also possible toinsert the pilot symbols as “pseudo-subchannels” after the DFT operationinstead of as time domain symbols before the DFT.

After block assembly, the blocks of mapped symbols (each a vector of (I,Q) constellation points) are typically supplied to each of M transmitchannel equalizers 2308 (“Tx-CE-m” for m=1 to M in FIG. 23A). EachTx-CE-m 2308 applies K blocks of amplitude and/or phase weights (eachamplitude phase weight may be represented in Cartesian form as a“complex” weight with a “real” and “imaginary” sub-component values)respectively to each of the K blocks of mapped vector symbols.

An exemplary embodiment of a transmit channel equalizer Tx-CE-m 2308 isshown in FIG. 24. The blocks of mapped vector symbols are {right arrowover (TxBlock)}_(k) wherein each symbol element within the vector {rightarrow over (TxBlock)}_(k) has an (I, Q) or, relative to FIG. 24,“complex” value (I representing a “real” sub-component value, Qrepresenting an “imaginary” sub-component value). Each {right arrow over(TxBlock)}_(k) vector of complex symbols is then complex multiplied by arespective transmit weight vector {right arrow over (WT)}_(m,k) as shownin FIG. 24. The channel equalized vector of symbols for each transmitchain m is then {right arrow over (TxBlockEq)}_(m) (the transmit chainchannel-equalized symbols) which represents the complex summation of thesymbol by symbol output vectors of each respective weighting operation.Note that FIG. 24 depicts an exemplary implementation of transmitchannel equalizer TX-CE-m 2308 based on vector complex multiplicationand summation. Such an approach is easily implemented using eithergeneric function calls or custom software modules executing on anembedded processor (or Digital Signal Processor—“DSP”). However,numerous other implementation techniques are also known such asdedicated logic circuits for complex multiplication and/or summation, aset of four scalar multiplier circuits and two scalar adders, or otherlogic circuits such as combinatorial gates (i.e. OR, NOR, AND, NAND,XOR, XNOR, etc.), multiplexers, shift registers, etc. that can producean equivalent result.

Note further that for embodiments with either OFDM or SC-FDE modulationand no frequency selective channel equalization, {right arrow over(WT)}_(m,k) would typically be composed of a block of identical transmitweights applied equally to all mapped symbols for a given stream k. Insome OFDM or SC-FDE embodiments where K=M, no equalization or weightingamongst transmit streams and chains may be desired such that eachTx-CE-m 2308 in FIG. 23A (or FIG. 24) is equivalent to a throughconnection mapping each {right arrow over (TxBlock)}_(k) directly toeach {right arrow over (TxBlockEq)}_(m) for each k=m.

For each transmit chain Tx-m, the channel equalized vector of symbolsfor each transmit chain {right arrow over (TxBlockEq)}_(m) is suppliedto Tx-Mux-m 2312, where m=1 to M, as shown in FIG. 23A. In the case ofOFDM modulation or SC-FDE modulation with frequency-selective channelequalization, each TX-Mux-m 2312 includes an IDFT to transform thesuccessive frequency domain symbols into a block of time domain samples.For SC-FDE modulation with no frequency selective channel equalization,and hence no DFT pre-coding within each Tx Block Assembler k 2304, then{right arrow over (TxBlockEq)}_(m) is already effectively a block oftransmit chain time domain samples and no such IDFT is required.

With reference back to FIG. 23A, each respective Tx-Mux-m 2312 isfollowed by a CPA and Block Serializer-m 2316. “CPA” refers to “CyclicPrefix Addition”—cyclic prefix was described above—which is performed bythe cyclic prefix adder within each CPA and Block Serializer-m 2316.Each CPA and Block Serializer-m 2316, via a cyclic prefix adder,typically replicates a set of samples from the end of the block andprepends these samples to the beginning of a cyclically-extended blockor includes logic to read out these samples in a way that produces anequivalent result. The CPA and Block Serializer-m 2316, via a blockserializer, then effectively performs a parallel to serial conversion ofthe extended block into a sequence of cyclically-extended (I, Q)transmit chain time domain samples.

The cyclically-extended block of (I, Q) time domain samples for eachtransmit chain Tx-m are then supplied to a respective Tx-DFE-m 2320 (DFErefers to Digital Front End). Each Tx-DFE-m 2320 performs a variety ofpossible time domain signal processing operations that may be specificto particular IBR embodiments. Common DFE processes in the transmit pathinclude digital up-conversion of the sampling rate, DC offsetcalibration for each Tx-m transmit chain, digital pre-distortion toaccount for non-linearities in the analog transmit path, pulse-shapingfiltering, crest factor or peak to average power ratio (PAPR) reduction,or frequency shifting to an “intermediate Frequency” (IF) suitable foranalog up conversion to RF-Tx-m in the Tx-m transmit chain (as opposedto the baseband (I_(T), Q_(T))_(m) interface illustrated in FIG. 16).Except for the last listed option of digital IF up conversion, theoutput (a transmit chain input signal) of each respective Tx-DFE-m 2320is typically a sequence of cyclically-extended blocks of calibrated,compensated and filtered baseband symbol samples (I_(T), Q_(T))_(m).

In the receive path of the exemplary IBR Channel MUX 628, downconvertedand amplified samples from I and Q ADCs in each Rx-n receive chain (areceive chain output signal) are passed to respective Rx-DFE-n 2340.Although the receive path of the IBR generally follows the logic ofreversing the operations of the transmit path, the details areconsiderably different in the IBR Channel MUX because in the receivepath the samples are corrupted by channel propagation impairments andarriving at initially unknown times. In view of this the receive pathDigital Front Ends of the IBR Channel MUX shown in FIG. 23B, Rx-DFE-n2340, perform several time domain digital operations typically includingmatched filtering, rotating, sampling rate downconversion, and Rx-nchain calibration. At minimum, the Rx-DFE-n 2340 is used in thedetection of timing synchronization information in the optional Preambleof the L PPDU-l transmissions from the transmitting IBR.

FIG. 26 illustrates this Preamble in the time domain. Note that theoptional Preamble for the l-th Tx-path in a transmitting IBR could beeither generated from a symbol library (as indicated optionally in Pilot(And Preamble) Symbol Library 2328 of FIG. 23) or read directly from aPreamble Samples Library 2324 (also indicated optionally in FIG. 23).

Referring again to FIG. 23B, the Acquisition/Synchronization detector2336 acquires a timing reference, for example, from detection of one ormore of the up to L optional preambles, one or more of the blocks ofTraining Pilots, an optional known pattern within a PLCP header, and/orpilot symbols within blocks of transmit symbol streams. TheAcquisition/Synchronization detector 2336 provides a feedback signal toeach Rx-DFE-n 2340 that enables, for example, symbol rotation andalignment in the time domain (typically in the RX-DFE-n using a CORDICalgorithm). The Acquisition/Synchronization detector 2336 furtherprovides symbol boundary and block boundary timing references (notshown) for use by various elements of the IBR Channel MUX 628 and theIBR Modem 624.

Such corrected receive symbol samples are then supplied to a respectiveCPR and Block Assembler-n 2344. “CPR” means “cyclic prefix removal.” TheCPR and Block Assembler-n 2344 effectively discards the number ofreceived samples at beginning of each block corresponding to the numberof cyclic prefix samples prepended to each block in the transmit path ofthe transmitting IBR. The remaining samples are serial to parallelbuffered into a single block (or an equivalent operation) suitable fordecomposition into receive chain frequency domain subchannels in eachrespective Complex DFT-n 2348.

As shown in FIG. 23B, the sets of receive chain frequency domainsubchannel samples output from the N Complex DFT-n 2348 are collectivelysupplied to L frequency domain receive channel equalizers 2352, each aRx-CE-l. FIG. 25 illustrates an exemplary embodiment of Rx-CE-l 2352.The receive channel equalizer Rx-CE-l 2352 is analogous to the transmitchannel equalizer Tx-CE-m 2308 shown in FIG. 24. Although implementationdetails of each Rx-CE-l 2352 can vary considerably just as describedabove for TX-CE-m 2308, essentially the frequency domain receive vectorsof I and Q samples for each block, {right arrow over (RxBlock)}_(n), arerespectively complex multiplied by complex receive weights {right arrowover (WR)}_(l,n) and then complex summed to produce a set ofchannel-equalized frequency domain estimates, {right arrow over(RxBlockEq)}_(l), representative of the original transmitted blockcorresponding to stream l.

The task of producing such complex receive weight vectors {right arrowover (WR)}_(l,n) that allow each Rx stream l to be separated from themyriad of signals, both desired and undesired, received via N (N≥L)receive chains is performed by the Channel Equalizer CoefficientsGenerator 2332 of FIG. 23A. There are many known algorithms forgenerating the appropriate frequency selective complex receive weightvectors for the IBR Channel MUX as shown in FIG. 23 wherein a known“Training Block” of pilot symbols for each stream l is used to determinethe receive weight vectors {right arrow over (WR)}_(l,n) that allowlinear detection of each {right arrow over (RxBlockEq)}_(l) vector inview of the actual receive chain frequency domain subchannel samplesobserved for the N receive chains during a reception periodcorresponding to transmission of the Training Block (or block ofTraining Pilots). In general, a Channel Equalizer Coefficients Generator2332 that is based on these algorithms calculates the various weights bycomparing the actual receive chain frequency domain subchannel samplesobserved during the reception of the Training Block with the expectedfrequency domain subchannel samples corresponding to a particular knownTraining Block for any given transmit stream. Such algorithms includeZero Forcing (ZF), Maximal Ratio Combining (MRC) and Minimum Mean SquareError (MMSE). Alternatively, other known algorithms for non-lineardetection of each {right arrow over (RxBlockEq)}_(l) are also known,such as Successive Interference Cancellation (SIC) in combination withZF, MRC or MMSE, V-BLAST (an acronym for “vertical Bell Labs LayeredSpace-Time) with ZF or MMSE, or Maximal Likelihood (ML).

The Channel Equalizer Coefficients Generator 2332 also supplies thecomplex transmit weight vectors {right arrow over (WT)}_(m,k) used fortransmit channel equalization within the exemplary IBR Channel MUX 628.In an ideal PTP TDD configuration where K=L, M=N, and no otherco-channel interference beyond the multiple transmit streams, forfrequency domain transmit channel equalization, a straightforwardalternative is to derive {right arrow over (WT)}_(m,k) directly from thecomputed {right arrow over (WR)}_(l,n) of the previous superframe.However, this can be sub-optimal for situations in which a co-channelinterferer, such as for example either another IBR PTP or PMP link inthe same vicinity or a conventional PTP link nearby, affects thereceived signals at the N receive chains of the AE-IBR differently fromthose at the RE-IBR. An alternative is to calculate {right arrow over(WR)}_(l,n) using both MMSE (which will maximize SINR at the receiver)and MRC (which will maximize SNR, or effectively maximize signal power,at the receiver) and then derive {right arrow over (WT)}_(m,k) from thecomputed, but unused in receive channel equalization, {right arrow over(WR)}_(l,n) for MRC. Note that for SC-FDE with no frequency selectivechannel equalization, then the constant {right arrow over (WT)}_(m,k)values applied to all symbols in a {right arrow over (TxBlock)}_(k) canbe blended using known algorithms from vector {right arrow over(WT)}_(m,k) values derived for MRC. This allows such an SC-FDE PTPtransmitter to improve the overall signal quality at the other receiverwhile allowing the other receiver to equalize both interference andfrequency selective fading effects. In view of the foregoing, it isadvantageous for the PMP AE-IBR to use either OFDM or SC-FDE withtransmit frequency selective channel equalization such that the ChannelEqualizer Coefficients Generator can compute MIMO-SDMA complex weights,using for example known EBF or Space Time Adaptive Processing (STAP)algorithms, to minimize multi-stream interference to RE-IBRs thatbeneficially receive only a subset of the transmitted streams. If SC-FDEwithout frequency selective transmit equalization is used, there isstill a benefit in using scalar {right arrow over (WT)}_(m,k) transmitweights at the AE-IBR derived for streams to a given RE-IBR fromprevious superframe MRC at the AE-IBR receiver but some additionalsignal separation such as time and/or frequency may be required todirect data to specific RE-IBRs.

With reference back to FIG. 23B, each respective channel equalized andinterference cancelled symbol samples stream vector {right arrow over(RxBlockEq)}_(l) (or set of channel-equalized frequency domainestimates) is then supplied to an Rx-MUX-l 2356. For OFDM, this istypically a parallel to serial converter that removes the pilotsubchannel symbol samples which are not needed in the IBR Modem. ForSC-FDE, each Rx-MUX-l 2356 includes a Complex IDFT and a parallel toserial converter that removes the time domain symbol samples at thepilot symbol sequence positions. Note though that for IBR embodimentswhere the transmitter uses SC-FDE with frequency selective equalizationand inserts the pilot symbols as “pseudo-subchannels” (see above), eachRx-MUX-l also needs to discard the pilot subchannels prior to theComplex IDFT transformation back to a time domain symbol samples stream.The output of each Rx-MUX-l is a receive symbol stream l that isprovided to a respective input to a demodulator core j.

FIG. 26 illustrates block by block generation of a given transmit symbolstream l in the transmitter IBR as combined into a transmit chain m. Thedesignator “l” is used in FIG. 26 instead of “k” for the transmitter todenote the generation of what becomes the “l-th” receive symbol streambeing demodulated at the receiver. Per the above description of FIG. 23,optionally every PPDU-l can start with a Preamble of known samples thatenables rapid signal acquisition and coarse synchronization. However,for some embodiments of the IBR in either FDD or in TDD with fixed andshort superframe timing, it is not necessary to perform such acquisitionand synchronization from the start of every PPDU. Thus, overhead can beadvantageously minimized by either discarding the Preamble from thePPDU-l generation process, at least for subsequent PPDUs after startup,or effectively combining the desired characteristics of the Preambleinto the Training Block via the Pilots-l. This approach can also be usedwith a Channel Sense Multiple Access (CSMA) approach to MAC schedulingto the extent that a timing master in the system, usually the AE-IBR,maintains sufficiently frequent timing synchronization transmissions.

In some embodiments, the PLCP Header for stream-l is matched to aparticular block such that the modulation and coding scheme (MCS) ofsuch PLCP Header block (see, for example, Block 1 of FIG. 26) is alwayspredetermined and usually chosen to deliver the PLCP header with higherreliability than the data payload blocks to follow. In typicalembodiments, the MCS includes an index number that conveys informationregarding parameters used in the modulation mapping and the forwarderror correction encoding processes. In such embodiments, the PLCPHeader typically conveys at least the information needed to inform thereceiver at the receiving-end IBR of the MCS used for the data payloadblocks to follow. In other embodiments, the MCS or informationindicating an increment, decrement or no change amongst a list ofpossible MCS values is conveyed within Block 0 of FIG. 26 by, forexample, adjusting the modulation of certain pilot symbols (or pilotsubchannels). In the case where Block 1 of FIG. 26 can carry more datathan just the PLCP Header alone, then the IBRs may allocate PPDU payloadto Block 1 even though the bits may be transferred at a different MCSthan that of subsequent Blocks.

In some embodiments, each Block 1 through f (or Block 1 through r) maycorrespond to the output of a block encoder and/or block interleaver ofFIG. 19 or 21. “f” denotes a forward link transmitted from an AE-IBR toan RE-IBR and “r” denotes a reverse link transmitted from an RE-IBR toan AE-IBR. Furthermore, each Block may include error detection bits(such as an FCS) or error correction bits (such as for a Reed-Solomon ora Reed-Muller outer encoder) that can be used either for an AutomaticRepeat reQuest (ARQ) re-transmission protocol of certain blocks or forinput to an RLC and/or RRC entity at either the AE-IBR or RE-IBR tooptimize future link parameters or resource selections.

FIG. 27 illustrates an exemplary embodiment of the Tx PLCP generator1804 of the IBR modem of FIG. 18. As shown in FIG. 27, the Tx PLCPgenerator 1804 includes a Tx PLCP Demux 2704 and multiple Tx PLCPModulators 2708. Based on input from the RRC 660, the Tx PLCP Demux 2704provides data interface streams TxD-j 2716, for j=1 to Jmod, torespective TxPLCP Mod-j 2708 that provide transmit data interfacestreams Tx-j 2720 to a respective Modulator Core j 1812 (shown in FIG.18).

In a PTP IBR configuration, such a plurality of data interface streams2716 and Modulator Cores 1812 may be used, for example, to provide linkdiversity, such as different sets of RLC parameters, carrier frequenciesand/or antenna selections where each set has appropriately multiplexedstreams and chains, or to provide probing capability. In a PMP IBRconfiguration, in addition to the above for PTP, such a plurality ofdata interface streams and Modulator Cores may also be used at an AE-IBRto optimize data transfer to certain RE-IBRs (or groups of RE-MRs) byusing techniques such as SDMA.

FIG. 28 illustrates an exemplary embodiment of the TxPLCP Mod-j 2708 ofFIG. 27. As shown in FIG. 28, the TxPLCP Mod-j 2708 includes a PLCPheader generator 2804, a PLCP input buffer 2808, a PLCP controller 2812,a training data library 2816, a padding generator 2820 and a PLCPtransmit MUX 2824. In FIG. 28, the PLCP Controller 2812 takes input fromthe RLC and the RRC as well as the current state of the PLCP InputBuffer 2808 to perform “rate matching” for a given PPDU. Rate matchingtypically is used to determine a particular choice of modulation andcoding scheme (MCS) in view of the instantaneous data transfer demandand recent link conditions. In addition to choosing an MCS, the PLCPController 2812 supplies information to the PLCP Header Generator 2804and directs the PLCP Input Buffer 2808 and the Padding Generator 2820 tosupply data to the PLCP Transmit Mux 2824. Depending on the status ofthe RRC and the instantaneous data transfer demand, the PLCP Controller2812 also enables training data from the Training Data Library 2816 tobe multiplexed into a given PPDU as indicated by certain fields in thePLCP Header. Typically, in some embodiments, the PLCP Header conveys atleast the MCS of the PPDU after Block 1, the length of the PPDU payload(for example, either in bytes of payload or in number of blocks pluslength in bytes of padding), the MCS of any training data, the selectedtraining dataset, and the number of training data blocks. Optionally,reply acknowledgements (ACKs) or non-acknowledgements (NACKs) for one ormore previous receive PPDUs corresponding to a companion receive datainterface stream j may also be sent in the PLCP Header. To the extentthat the PLCP also concatenates multiple MPDUs and/or fragments thereof,the PLCP Header Generator also includes sufficient information to allowthe RxPLCP to recover such MPDUs and/or fragments upon receipt. Asdescribed above, FIG. 26 illustrates an exemplary PPDU having a PLCPHeader, a PPDU payload including one or more MPDUs or fragments thereof,and a PAD to cause the PPDU to occupy an integer number of Blocks asshown. Not shown in FIG. 26 is a PPDU extension with one or more Blocksof data from the Training Data Library 2816 of FIG. 28 as may beindicated by the PLCP Header.

Note that for Modulator Cores 1812 that divide each transmit datainterface stream Tx-j into two or more transmit symbol streams, the PLCPHeader of FIG. 28 may be an amalgam of two or more PLCP Headers forrespective transmit symbol streams especially in the case where aseparate MCS applies to each stream (the “vertical encoding” case shownfor example in FIG. 21). Alternatively, a single PCLP Header may be usedper Modulator Core j 1812 with parts of the PLCP Header apportionedamongst two or more transmit symbol streams (the “horizontal encoding”case shown for example in FIG. 19). Another alternative would have asingle PLCP Header encoded and replicated across all transmit symbolstreams of a particular Modulator Core j 1812. In some embodiments, eachPLCP Header further includes an FCS (not shown in FIG. 28) to enable theTx PLCP to determine if the PLCP Header has been demodulated and decodedwithout error. Note also that the PLCP Controller 2812 may be used incertain embodiments to provide or control the provision of timingsignals to the IBR Channel MUX 628 (not shown in FIG. 23) that enableinsertion of either or both of the optional Preamble or of the block ofTraining Pilots as shown in FIG. 26. In exemplary embodiments, a blockof Training Pilots, used, for example, to update transmit and/or receiveweights by the IBR at an opposite end of a link, may be inserted at thebeginning of every transmit superframe, the beginning of every PPDU (asshown in FIG. 26), or at such intervals within a PPDU or superframe asmay be communicated to the Tx PLCP generator 1804 or IBR Channel MUX 628via the RRC 660.

FIG. 29 illustrates an exemplary embodiment of the Rx PLCP analyzer 1808of FIG. 18. As shown in FIG. 29, the Rx PLCP analyzer 1808 includes a RxPLCP MUX 2904 and multiple Rx PLCP Demod-j 2908. In FIG. 29, multiplereceive data interface streams Rx-j each from respective DemodulatorCore j 1812 (see FIG. 18) are input to a respective Rx PLCP Demod-j2908. After removing the respective PLCP Headers, Padding (if any), andTraining Data (if any), each RxD-j is then multiplexed in the RX PLCPMux 2904 and transferred to the IBR MAC 612 as Rx Data 620.

FIG. 30 illustrates an exemplary embodiment of the Rx PLCP Demod-j 2908of FIG. 29. As shown in FIG. 30, the RxPLCP Demod-j 2908 includes a PLCPheader analyzer 3004, a PLCP output buffer 3008, a PLCP controller 3012,a training data analyzer 3016, and a PLCP receive Demux 3024. The PLCPController 3012 may be shared in some embodiments with the PLCPController 2812 of FIG. 28. The Tx or Rx PLCP Controllers 2812, 3012 mayalso be shared across some or all data interface streams. When a PPDU isreceived from a receive data interface stream Rx-j, the PLCP HeaderAnalyzer 3004 filters certain fields from the PLCP Header and sends thisinformation to the PLCP Controller 3012 (or alternatively, is a part ofthe PLCP Controller 3012). This enables the exemplary PLCP Controller3012 to signal the MCS for subsequent symbols or blocks to DemodulatorCore j 1812 as shown in FIG. 30 and to control the PLCP Receive Demux3024 such that the receive data interface stream Rx-j is directed to thePLCP Output Buffer 3008 or Training Data Analyzer 3016 or ignored in thecase of Padding. Further information from the PLCP Header Analyzer 3004and/or PLCP Controller 3012 enables the Training Data Analyzer 3016 todetermine the match between transmitted and received/demodulated/decodedtraining data and communicate certain metrics such as the number of biterrors and the distribution of errors to the RRC 660. In embodimentswhere the PLCP header carries an FCS, the PLCP Header Analyzer 3004 alsochecks the FCS (not shown in FIG. 30) and if a failure occurs, signalsat least the PLCP Controller 3012, the RRC 660, and the RLC 656. ThePLCP Controller 3012 may optionally also inhibit sending RxD-j for theinstant PPDU in the case of such an FCS failure or it may send it onwith an indicator (not shown) to the IBR MAC that such an FCS failurehas occurred.

With reference back to FIGS. 6 and 7, other IBR elements include the IBRMAC 612, the Radio Link Control (RLC) 656, the Radio Resource Control(RRC) 660 and, specific to FIG. 7, the IBMS Agent 700. Although IBRembodiments are possible wherein the MAC 612, RLC 656, RRC 660 and IBMS700 are distinct structural entities, more commonly IBRs are realizedwherein the IBR MAC 612, RLC 656, RRC 660, IBMS 700 and portions of theIBR Interface Bridge 608 are software modules executing on one or moremicroprocessors. Note also that in some IBR embodiments that use of a“Software Defined Radio” (SDR) as the IBR Modem 624 and/or IBR ChannelMUX 628 or portions thereof may also be realized in software executingon one or more microprocessors. Typically in SDR embodiments, the one ormore microprocessors used for elements of the PHY layer are physicallyseparate from those used for the MAC 612 or other layers and arephysically connected or connectable to certain hardware cores such asFFTs, Viterbi decoders, DFEs, etc.

FIGS. 31 and 32 illustrate exemplary views of the IBRs of FIGS. 6 and 7,respectively, showing exemplary communications protocols stacks. It willbe appreciated that some embodiments have the one or more 802.1 MACinstances, the IBR LLC, the IBR MAC, the RRC, the RLC, the IBMS and allother upper layer protocols implemented as software modules or processesexecuting on one or more microprocessors within the IBR.

In accordance with the previous description of the IBR Interface Bridge608, the 802.1 MAC instances of FIGS. 31 and 32 correspond respectivelyto Ethernet interfaces 604 and are substantially conventional. Each802.1 MAC instance is paired with a respective 802.3 PHY instance ofwhich exemplary types include 100 Base-T, 1000 Base-T and variousmembers of the 1000 Base-X family of fiber optic interfaces.

In many exemplary embodiments of the IBR, the IBR Logical Link Control(IBR LLC) layer of FIGS. 31 and 32 is compatible with IEEE 802.2. Inaccordance with the previous description of the IBR Interface Bridge,the IBR LLC layer also uses substantially conventional processes tobridge, for example, 802.2 compatible frames across disparate mediatypes as exemplified by the 802.1, the 802.11 and the IBR MAC layers ofFIGS. 31 and 32. Furthermore, in accordance with the previousdescription of the IBR Interface Bridge, the IBR LLC layer also usessubstantially conventional processes to switch and/or load balance 802.2compatible frames amongst multiple 802.1 MAC instances if available.

FIGS. 31 and 32 depict an exemplary IEEE 802.11 compatible radio (alsoknown as “WiFi”) including 802.11 MAC and 802.11 PHY layers and one ormore antennas. In some IBR embodiments, such an 802.11 radio, which maybe compatible with 802.11b, 802.11a, 802.11g, 802.11n or other 802.11PHY layer variants, within the IBR may be configured as an access point(AP) to allow significant wireless local area network (WLAN) traffic tobe backhauled by the IBR over its wireline or wireless interfaces asappropriate. In some embodiments, the same antennas may be utilized inboth the IBR PHY, which typically includes the IBR Modem 624, IBRChannel MUX 628, IBR RF 632 and IBR Antenna Array 648 in reference toFIGS. 6 and 7, and the 802.11 PHY (e.g., via the IBR RF Switch Fabric ofFIG. 10 (not shown)). Note further that multiple instances of the 802.11radio can also be connected to the IBR LLC in an analogous manner toconnecting multiple Ethernet interfaces.

Another IBR embodiment may provide a more limited 802.11 radiocapability for local configuration purposes (to the exclusion of WLANtraffic and general access). Traditionally PTP and PMP systems haveprovided a “console” input for local configuration via a command lineinterface of radios in the field particularly for situations wherenetwork configuration is either unavailable or undesirable. However,when an IBR is deployed, for example, on a street light, traffic light,building side, or rooftop, a wired console access with a terminal may beextremely inconvenient and/or dangerous. Thus, in some embodiments,802.11 radios are deployed as an access point such that terminalsincluding smartphones, tablets, laptops or other portable computingdevices with 802.11 station capability can connect to such IBRs in sucha console mode. In one embodiment, this deployment is used solely forconfiguration purposes. Configuration by a terminal with 802.11 stationcapability is also possible in the case of one or more 802.11 APsdeployed for WLAN access purposes. Note further that for theconfiguration-only 802.11 AP in an IBR, exemplary embodiments may alsodeploy the 802.11 radio as an Ethernet device on one of the P Ethernetinterfaces depicted in FIGS. 6, 7, 31 and 32, or via some otherIBR-internal wired or bus interface. In such configuration-only 802.11AP IBR embodiments, one or more separate antennas for 802.11 use wouldtypically be provided by the IBR. Furthermore, the configuration-only802.11 interface is not restricted to AP mode only but can also includepeer to peer (“IBSS”), AP mode at the terminal, or WiFi direct. In oneembodiment, the IBRs described herein can be used as fixed wirelessaccess radios or fixed wireless broadband access radios. These radiosare typically connected to, for example, homes, buildings, and the like.

With reference to FIGS. 31 and 32, typical applications protocols suchas Hyper Text Transfer Protocol (HTTP), Simple Network ManagementProtocol (SNMP), and others can also be implemented using substantiallyconventional software processes or modules executing on one or moremicroprocessors within exemplary IBRs. Such applications protocols canget access to or be accessed from any of the depicted network interfacesin FIGS. 31 and 32 via industry-standard Transmission Control Protocol(TCP) or User Datagram Protocol (UDP) transport layer protocols and theInternet Protocol (IP) network layer protocol. TCP, UDP and IP can beimplemented using substantially conventional software processes ormodules executing on one or more microprocessors within exemplary IBRs.Note that in certain deployments, such as backhaul within a cellularoperator's radio access network (RAN), messages to or from HTTP, SNMP orother applications protocols may need to be sent either via othertransport and network protocols or via encapsulation of TCP/UDP and IPsegments and packets within such other transport and network protocols(not shown in FIGS. 31 and 32). Note further that messages to or fromapplications protocols in the IBR may also bypass the TCP/UDP and IPprotocols, or their equivalents, entirely during a console mode (notshown) using a locally connected terminal.

FIGS. 31 and 32 also illustrate an exemplary element entitled “IBRControl” which is typically implemented as a distribution plane ofinterconnects, buses and/or controllers that arbitrate directcommunications between the RRC and RLC entities with various elements ofthe IBR MAC and IBR PHY. Because the IBR Control does not passinformation amongst such entities by using the communications protocolsstack, such information can be communicated with minimal latency forreal time control.

FIG. 33 illustrates an exemplary implementation of the IBR MAC 612. TheIBR MAC 612 is shown in terms of various functional elements and some oftheir relationships to each other as well as to the IBR PHY, the IBRLLC, the RRC, the RLC and the IBR Control. Operation of the IBR MAC ofFIG. 33 may depend on the type of “superframe” timing utilized.Typically, IBRs send a single PPDU per Modulator Core-j (or per transmitstream k), which may include Training Data blocks as described above, ina given transmit superframe (TxSF) and conversely will receive a singlePPDU per Demodulator Core-j (or per receive stream l) in a given receivesuperframe (RxSF).

As shown in FIG. 33, the IBR MAC 612 includes a IBR MAC managemententity 3304, a MAC Tx Buffer and Scheduler 3308, an IBR MAC ControlEntity 3312, a MAC Rx Buffer 3316, a decryption block 3320, anencryption block 3324, a MPDU header generator 3328, a first FCSgenerator 3332, a MPDU header analyzer 3336, a second FCS generator 3340and a FCS Pass? analyzer 3344.

FIG. 34 illustrates channel activity for a modulator and demodulatorpair j with data interface streams Tx-j and Rx-j respectively versustime for an exemplary IBR using FDD with fixed superframe timing.Similarly, FIG. 35 illustrates analogous channel activity for TDD withfixed superframe timing and FIG. 36 illustrates analogous channelactivity for TDD and Collision Sense Multiple Access (CSMA) withvariable superframe timing.

With reference back to FIG. 33, the operation of the IBR MAC is firstdescribed in the context of an example based on TDD/CSMA with variablesuperframe timing. As shown in FIG. 36, in CSMA transmissions by eitherthe AE-IBR or RE-IBR paired in an active link, whether PTP or PMP,defers to other channel activity detected within its present receiveantenna pattern. In practice, this can be achieved by using the receiveportion of the IBR PHY to determine in channel and in-view signal energywhich if above a threshold causes an inhibit control signal at either aTx PLCP Mod-j of FIG. 27 (not shown) corresponding to the affected linkor at the MAC Tx Buffer and Scheduler of FIG. 33 (not shown) identifyingthe affected link. For purposes of this description, the latter optionis considered. Furthermore, it is assumed that for this specific exampleevery MSDU or frame request at the MAC Tx Buffer and Scheduler resultsin a single MPDU based on such MSDU or frame request (see, for example,FIG. 4) that is then matched to a single PPDU in the Tx PLCPcorresponding to a given link. Other variants are possible with the IBRMAC and IBR PHY as discussed subsequently.

With reference again to FIG. 33, the primary source of MSDUs to the MACTx Buffer and Scheduler 3308 is typically in the data plane from the IBRLLC 3348. Either MSDUs and/or frame requests in the control plane canoriginate from the RRC 660 (or indirectly from the optional IBMS Agent700), the RLC 656, the IBR MAC Management Entity 3304, or the IBR MACControl Entity 3312. Exemplary details of frame requests and/or MSDUsoriginating from the RRC 660 or RLC 656 are discussed subsequently.

Exemplary frames originating at the IBR MAC Management Entity 3304include those associated with management processes such as Association,Authentication and Synchronization. In some cases, the IBR MACManagement Entity 3304 may send an MSDU payload to the MAC Tx Buffer andScheduler 3308 with a specific frame request type. In other cases, theIBR MAC Management Entity 3304 may send only the specific frame requesttype wherein all relevant information to be conveyed to the receivingIBR(s) will be present in the MPDU Header to be generated based on thedetails of the frame request type and other information presently knownto the IBR MAC.

In some embodiments, Association and Authentication processes can occurvia exchange of management frames in a substantially conventionalfashion. A particular RE-IBR may choose to associate with a given AE-IBRby sending an Association Request management frame directed to theAE-IBR based on either advertised information received from such AE-IBRand/or configuration information currently present in the RE-IBR. Uponreceipt of an Association Request, the AE-IBR can proceed according toits presently configured policies to associate with or deny associationto the RE-IBR. If associated, the AE and RE would exchangeauthentication frames in substantially conventional fashion to ensurethat compatible encryption/decryption keys are present for subsequentframe exchanges.

In exemplary IBRs, an RE-IBR can, for example, associate with adifferent AE-IBR if its present AE-IBR (or its wireline link interface)fails, the link throughput falls below a minimum threshold, the presentAE-IBR forces a disassociation, the present AE-IBR inhibits linkresource allocations to the RE-IBR below a minimum threshold, or theRE-IBR becomes aware of a preferred AE-IBR, all as set by certainconfiguration policies at the time as may be set by the optional IBMSAgent 700 as shown in FIG. 33. Furthermore, certain exemplary IBRs witha plurality of modulator cores can as an RE-IBR maintain a plurality ofcurrent associations with multiple AE-IBRs per configuration policies toenable enhanced throughput and/or link diversity.

Another set of exemplary management frames issued by the IBR MACManagement Entity 3304 concerns synchronization, status and presence.Periodically, (as configured or directed by the optional IBMS Agent 700)an exemplary AE-IBR may choose to send a Synchronization Frame thatadvertises certain capabilities of the AE-IBR including wireline linkfailure conditions and provides a time stamp reference usable byexemplary RE-IBRs for timing synchronization as either broadcastuniformly across the full directionality possible for the IBR AntennaArray and/or across all current links. Advantageously, particularly foran AE-IBR with multiple associated RE-IBRs in a PMP configuration, sucha Synchronization Frame (or other such management frame) can direct oneor more RE-IBRs to make internal reference timing offsets such that thetime of arrival of transmissions from such RE-IBRs is more optimallyaligned for best simultaneous reception at the AE-IBR in either FDD orTDD with fixed superframe timing (see FIGS. 34 and 35). For example, theAE-IBR may determine such timing offsets within theAcquisition/Synchronization element of the IBR Channel MUX 628 byanalyzing preamble samples corresponding to particular RE-IBRtransmissions.

With reference again to FIG. 33, exemplary frames originating at the IBRMAC Control Entity 3312 include those associated with control processessuch as Acknowledgement and Access Control. Analogously to the IBR MACManagement Entity 3304, the IBR MAC Control Entity 3312 may send an MSDUpayload to the MAC Tx Buffer and Scheduler 3308 and/or a specific framerequest type wherein relevant information can be conveyed to thereceiving IBR in the MPDU Header.

In exemplary IBRs, Acknowledgement Frames can provide an ACK or NACKindication for configured frame types recently received. The frames areidentified uniquely and then set to ACK or NACK based on the receivepath FCS comparison process illustrated in FIG. 33. To minimizetransport overhead, implementation complexity and transmit latency, someembodiments of the IBR utilize a NACK protocol administered by the IBRMAC Control Entity. With reference to FIGS. 34-36, upon receipt of oneor more MPDUs within RxSF(s), where “s” is a superframe sequence number,then if any MPDU has an FCS failure within RxSF(s), a NACK bit is set inthe MPDU Header of the first (or each) MPDU within TxSF(s+1). If theoriginating IBR detects a positive NACK in TxSF(s+1) at its receiver,then such an IBR will re-transmit via its MAC Tx Buffer and Scheduler3308 the MPDU (or MPDUs) associated with RxSF(s) using link parametersas determined by its RLC 656 at that time. For the FDD case of FIG. 34,such re-transmitted MPDU (or MPDUs) originally from RxSF(s) would atleast partially be received in RxSF(s+2) whereas for the TDD cases ofFIGS. 35 and 36, it would be RxSF(s+1). If there were no FCS failures inRxSF(s), or the originating IBR fails to detect a positive NACK inTxSF(s+1) for any reason, then the originating IBR will clear the MPDU(or MPDUs) associated with RxSF(s) from its MAC Tx Buffer and Scheduler3308. Note that unlike conventional ACK protocols, this exemplary IBRNACK protocol does not guarantee 100% delivery of MPDU. However,occasional MPDU delivery failures are typically correctable at a higherlayer for data plane MSDUs (i.e., using TCP or an equivalent transportlayer protocol), non-catastrophic for management or control plane framesto the IBR MAC Management Entity 3304, the IBR MAC Control Entity 3312,the RRC 660, or the RLC 656, and preferable to the increased latency andjitter of conventional ACK protocols applied to backhaul for those dataplane MSDUs not subject to higher layer correction (i.e. using UDP or anequivalent transport layer protocol). Note further that this NACKprotocol is also advantageously used when PLCP Header FCS failures occurby, for example, having the Rx PLCP Demod-j discarding such a failedPPDU but substituting an MPDU with correct link ID but dummy FCS toRxD-j in FIGS. 29 and 30 to trigger an FCS failure at the IBR MAC 612.

In some embodiments, Access Control Frames are initiated at the IBR MACControl Entity 3312 to control the behavior of other IBRs with currentlinks to the initiating IBR. Such Access Control Frames can, forexample, restrict the rate at which data plane MSDUs are sent, restrictthe timing in which data plane MSDUs are sent, or temporarily inhibitfurther data plane MSDUs from being sent in a local overloadingscenario. For example, an exemplary AE-IBR could utilize Access ControlFrames to enforce access policies discriminatorily amongst multipleRE-IBRs according to certain configuration parameters such as a ServiceLevel Agreement (SLA). Such access policies may also be set via theoptional IBMS Agent 700 as shown in FIG. 33.

With reference to FIG. 33, the MAC Tx Buffer and Scheduler 3308temporarily stores data plane MSDUs from the IBR LLC 3348 as well asframes or frame requests from the IBR MAC Management Entity 3304, IBRMAC Control Entity 3308, RRC 660 and/or RLC 656 until such data can bescheduled for transmission within one or more MPDUs. Such scheduling isadvantageously performed based on policies either configured locally oroptionally communicated from the IBMS Agent 700 via the RRC 660 and theIBR Control 3352. In some embodiments, data plane MSDUs from the IBR LLC3348 usually are the lowest priority in the queue for schedulingcomposition of the next MPDU. If the rate of MSDU delivery from the IBRLLC 3348 exceeds the instant link delivery capability, then to avoidbuffer overflow the MAC Tx Buffer and Scheduler 3308 may have an Xon/offcontrol feedback to the IBR LLC 3348 to cause the IBR LLC 3348 to stopsending MSDUs until the appropriate buffer status returns. This mayresult in the IBR LLC 3348 causing certain other interfaces such as anEthernet or WiFi interface to reduce MSDU traffic. To the extent thatthe ratio of control plane traffic to data plane traffic is small, as isby design for some embodiments, then scheduling priority amongst framesother than MSDUs from the IBR LLC 3348 is unimportant and “first-in,first-out” is sufficient.

Note that the foregoing embodiment of the MAC Tx Buffer and Scheduler3308 performs scheduling based on frame type without regard for otherinformation such as the PCP within an 802.1 MAC frame as may be presentin MSDUs from the IBR LLC 3348. The IBR LLC 3348 in its bridgingcapacity may forward MSDUs to the MAC Tx Buffer and Scheduler 3308 inorder based on PCP (or other such QoS or CoS indicators) so that the MACTx Buffer and Scheduler 3308 need not repeat the queuing exercise. Inalternative embodiments, such QoS prioritization of MSDUs can also beperformed at the MAC Tx Buffer and Scheduler 3308 instead.

Upon scheduling an MPDU for transmission as described above, the MAC TxBuffer and Scheduler 3308 causes the MPDU Header Generator 3328 tocompose an MPDU Header specific to the pending MPDU. In conventionalIEEE 802 based communications systems, such an MPDU Header would includeat least the physical address (also known as the “MAC address” or the“hardware address”) of both the origination and destination IBRs.However, sending such physical addresses (typically 48 or, morerecently, 64 bits) in every MPDU unduly burdens the IBR links withunnecessary overhead that reduces MAC efficiency. Thus, in someembodiments, a Link Identifier (LID) is substituted in every MPDU headerinstead. Exemplary LID implementations can be as few as 16 bits inlength. For example, each exemplary LID may include an AE-IBR identifierof 10 bits and an identifier of 6 bits for an RE-IBR presentlyassociated with the AE-IBR. This is possible because in some embodimentsthe IBRs are configured in view of their fixed geographic positions inthe field as set at time of deployment or optionally controlled by theIBMS Agent 700 such that no overlapping AE-IBR identifiers are withinradio range of RE-IBRs possibly associated with them. The AE-IBR mayassign a locally unique RE-IBR association identifier field as part ofthe association process. For unassigned links, all zeros (or ones) canbe sent for LID and then the frame payload, for example a managementframe used in the association process, can include the full physicaladdresses as appropriate. Note that even if, in the alternative, thatlonger (possibly 24 bits or 32 bits) “regionally-unique” or even“globally-unique” LIDs were used, then because the overall number ofworldwide backhaul links is generally much less the overall number ofworldwide network devices, such extended length LIDs can still be muchshorter than traditional IEEE 802 based addressing schemes.

A Frame Type Identifier (FTI) may be placed in the MPDU header by theMPDU Header Generator 3328. In one embodiment, the FTI is no more than 5bits total. In one particular embodiment, the FTI is part of a FrameControl Field (FCF) of 8 bits, and the other 3 bits include 1 bit forthe NACK protocol control bit indicator (set to 1 if the previousRxSF(s) had an MPDU with an FCS failure), 1 bit to indicate if theinstant frame of the MPDU payload is encrypted, and 1 bit to indicate ifthe instant frame of the MPDU payload is the last frame (LF) in thepayload. Alternatively, the 1 bit for the NACK can be 1 bit for the ACKindicator (set to 1 if the previous RxSF(s) had an MPDU without an FCSfailure) if an ACK protocol is used for the instant MPDU FTI. Followingthis FCF byte, a 16 bit Fragment and Length Indicator (FLI) is placedsequentially in the MPDU header, wherein, for example, 3 bits of the FLIindicates by 1 bit if the instant frame payload is the last fragment andby 2 bits the fragment sequence number and 13 bits indicate the instantframe payload length in bytes. Following the 2 FLI bytes, an 8 bit FrameSequence Number (FSN) is placed sequentially in the MPDU header. The FSNare typically sequentially generated except where repeated for thoseframe payloads sent as fragments. If LF=1 in the initial FCF byte of theMPDU header (as would be the case for the single MSDU of frame payloadper MPDU scenario described above for TDD/CSMA), then the MPDU header iscomplete. If the MAC Tx Buffer and Scheduler 3308 is configured topermit concatenation of MSDUs or other frame payloads up to some maximumMPDU payload (as would be compatible with many TDD/CSMA deployments),then LF=0 when FLI describes an instant frame payload lengthsufficiently low to allow another available frame payload to beconcatenated within the maximum MPDU payload length and an additionalFCF, FLI and FSN combination would be generated at the MPDU HeaderGenerator 3328 and repeated until an FCF with LF=1 is encountered. Notethis process of concatenated FCF, FLI and FSN fields within an MPDUheader corresponding to concatenated frame payload can also beadvantageously applied to fixed superframe timing in either FDD or TDDas illustrated in FIGS. 34 and 35.

The MAC Tx Buffer and Scheduler 3308 further provides the one or moreframe payloads that form the MPDU payload to the Encryption element 3324to be encrypted on a frame payload by frame payload basis as indicatedin the FCF using substantially conventional encryption algorithms. Anexemplary algorithm suitable for the IBR is the Advanced EncryptionStandard (AES) which has been published by the National Institute ofStandards and Technology. In one embodiment, the IBRs use AES with a 256bit key length. Other key lengths and other encryption algorithms arealso possible for exemplary IBRs. Exemplary IBRs can also employencryption for all frames after encryption keys are exchanged duringauthentication (and even including association and authentication framesto the extent encryption keys sufficient at least for association andauthentication are provided to IBRs via, for example, factory setting orconsole mode interface).

The encrypted (and/or unencrypted as desired) frame payload(s) and theMPDU header are then concatenated together as shown in FIG. 33 and thenpassed through an FCS Generator 3332 that generates an FCS (e.g., of atleast 32 bits in length). Alternatively, some IBR embodiments may alsoencrypt (and decrypt) MPDU headers. Other IBR embodiments may attainsome measure of MPDU header privacy (and PLCP header privacy) by settingscrambler and descrambler parameters (see FIGS. 19-22) derived from, forexample, the encryption keys and/or LID. The FCS is then appendedfollowing the MPDU header and frame payload(s) as also shown in FIG. 33to complete the composition of an MPDU to be sent to the IBR PHY 3356over the Tx Data interface.

Note also that for those frame types, particularly certain managementframes originating at the AE-IBR of a PMP configured deployment, thatare intended to be broadcast to all current links at an IBR, such framepayloads may be distributed to all such links in parallel at the MAC TxBuffer and Scheduler 3312 such that the same frame payload is providedto at least one MPDU corresponding to each current link. The IBRstypically generate very little broadcast traffic and most Ethernet orWiFi broadcast traffic on the other IBR interfaces is filtered at theIBR LLC 3348.

With reference to FIG. 33, the receive path of the exemplary IBR MACperforms essentially the reverse operations described above in thetransmit path. Starting with the IBR PHY 3356, after the Rx PLCPanalyzer 1808 (see FIG. 18) recovers an MPDU, it is passed via the RxData interface in FIG. 33 to a splitter 3360 within the IBR MAC. Thesplitter 3360 removes the trailing bits of the MPDU corresponding to theFCS, or RxFCS. The remainder of the MPDU minus RxFCS is passed throughan FCS Generator 3340, the result of which is compared at FCS Pass?analyzer 3344 to determine if it is identical to RxFCS. If it isidentical, then a known-good (barring the extraordinarily rare falsepositive FCS event) MPDU is received. The FCS Pass? status iscommunicated to the IBR MAC Control Entity 3312, the MAC Rx Buffer 3316,the MPDU Header Analyzer 3336, and the RRC 660 and RLC 656 (foranalytical purposes) as shown in FIG. 33. The MPDU Header Analyzer 3336receives from a second splitter the leading bits of the MPDUcorresponding to a minimum length MPDU header described above. If LF=1,then the remaining payload portion of the MPDU is passed to theDecryption element 3320 (or bypassed if appropriate) for decryption whenthe encryption indicator bit of the FCF is set. The MPDU Header Analyzer3336 may also verify (or cause to be verified) that the payload lengthcorresponds to that described in the FLI field of the MPDU Header andthat the LID is valid. Upon decryption (as appropriate), the MPDUpayload is passed to the MAC Rx Buffer 3316 which then forwards the MSDUor frame payload to the IBR MAC Management Entity 3304, the IBR MACControl Entity 3312, the IBR LLC 3348, the RRC 660 or the RLC 656 asappropriately directed by the MPDU Header Analyzer 3336 based on LID andFTI. The MPDU Header Analyzer 3336 may also directly signal the IBR MACControl Entity 3312 with the status of the NACK bit in the exemplaryMPDU header described above. In the event that the instant frame payloadis a fragment, as evident by the FLI field in the exemplary MPDU headerdescribed above, then the MPDU Header Analyzer 3336 instructs the MAC RxBuffer 3316 based on the FSN field to either store the fragment,concatenate the fragment to one or more other stored fragments, ordeliver the multiple fragments if the instant fragment has the lastfragment indicator within the FLI field as described above. Optionallyif a positive ACK protocol is used for 100% delivery of every MPDU(versus the superframe NACK protocol described above), then the MDPUHeader Analyzer 3336 also verifies the FSN for duplicated detection anddiscards (or causes to be discarded) such duplicate payloads.

In the event that the MPDU being received has multiple concatenatedMSDUs and/or frame payloads as described optionally above, then the MPDUHeader Analyzer 3336 interacts with the second splitter via the feedbacksignal shown to continue parsing off consecutive MDPU header bytes, forexample the repeated exemplary FCF, FLI and FSN fields, until an FCFwith LF=1 is encountered. The above described process for the MPDUHeader Analyzer 3336 directing the Decryption element 3320 and the MACRx Buffer 3316 to deliver the multiple payloads is then repeated eitherserially or in parallel as desired until all contents of the MPDUpayload have been resolved by the IBR MAC.

In the event that the FCS Pass? analyzer 3344 described above determinesan FCS failure then the receive path of the exemplary IBR MAC operatesdifferently depending on certain options that may be designed into theIBR or selected based on configuration policies. For the superframe NACKprotocol described above, the FCS failure is directly communicated tothe RRC 660 and RLC 656 (via the IBR Control 3352 of FIGS. 31-33) fortheir analytical purposes and is directly communicated to the IBR MACControl Entity 3312 so that TxSF(s+1) is sent with NACK=1 in theexemplary MPDU header described above. Optionally, the MPDU HeaderAnalyzer 3336 may attempt to determine if an apparently valid MPDUheader is present, and, if so, attempt to deliver certain MSDUs or framepayloads as described above for the case of a known-good MPDU. However,as directed by configuration policies, certain frame types may bediscarded. For example, in one embodiment, only MSDUs to the IBR LLC3348 that correspond to apparently valid MPDU header fields would bedelivered in an FCS failure situation, and even then only afterbuffering such MSDUs for one additional superframe period to determineif a duplicate FSN MSDU may be available due to the re-transmissionattempt associated with the superframe NACK protocol described above.Alternatively, in the case of an ACK protocol that guarantees 100%delivery of known-good MPDUs, then the entire MPDU payload is discarded.

For the types of data throughput rates and superframe lengths that arepractical, the single payload MPDU per superframe example describedabove for the TDD/CSMA example of FIG. 36 is not an optimum choice forthe fixed superframe timing examples for FDD in FIG. 34 or TDD in FIG.35. One alternative uses concatenated multi-payload frame MPDUs up tosome maximum length and then forwards such MPDUs to the Tx PLCP tocompose PPDUs as described previously. Another alternative, compatiblewith the superframe NACK protocol described above, would have the MAC TxBuffer and Scheduler 3308 utilize the same information from the RLC 656and the RRC 660 available to the Tx PLCP and thus compose a singlemulti-payload MPDU that will, with appropriate padding added in the TxPLCP to an integer block count, maximally occupy the available PPDU forthe pending superframe.

Note that for the TDD case depicted in FIG. 35 with fixed superframetiming, IBRs are capable of operating in such modes wherein thesuperframe timing in the forward link from AE to RE may be differentthan in the reverse link. For a PMP AE-IBR this typically requires thatall RE-IBRs adopt the same superframe timing parameters in associationwith this particular AE-IBR. Some RE-IBRs, in either PTP or PMPconfigurations, may be simultaneously associated with two or moreAE-IBRs to the extent such IBRs permit this policy and have sufficientradio resources to maintain such links. Given that multiple AE-IBRsassociated with such RE-IBRs can adopt different TDD fixed superframetiming parameters relative to each other, such timing parameters may becommunicated to all RE-IBRs via management frames originating in the IBRMAC Management Entity 3304. Similarly, to the extent an RE-IBR islimited in superframe timing access due to such multiple associations,such an RE-IBR can communicate such restrictions to the AE-IBRs viamanagement frames from its IBR MAC Management Entity 3304.

For PMP configurations, the AE-IBR can advantageously transmit to orreceive from multiple RE-IBRs simultaneously in a given radio channelusing SDMA as described above. To the extent that the number of RE-IBRsassociated with an AE-IBR exceeds either the AE-IBR's available SDMAresources or certain RE-IBRs are not spatially separable by the AE-IBR,then simultaneous transmissions to such RE-IBRs would require multipleradio channels which in practice is often either undesirable orimpractical. Another alternative for maintaining links to RE-IBRswherein such SDMA capabilities are being exceeded is through the use ofTime-Division Multiplexing (TDM). For either an FDD or TDD system withfixed superframe timing, such as depicted in FIGS. 34 and 35, an AE-IBRcan use TDM to designate certain “subframes” within a given superframeapplied to a modulator/demodulator resource pair j to particular LIDs.Management frames from the AE-IBR's IBR MAC Management Entity 3304 can,in exemplary embodiments, inform affected RE-IBRs of such LID to TDMsub-frame mappings and associated subframe timing parameters. In someembodiments, any RE-IBR associated with two or more AE-IBRssimultaneously could request that such LID to TDM sub-frame mappingsand/or timing parameters account for its local timing preferences foroptimally maintaining such simultaneous links. The MAC Tx Buffer andScheduler 3308 may compose MPDUs in view of TDM sub-frame timingparameters that are assigned such that the TxPLCP can optimally utilizean integer number of transmit blocks in each sub-frame. The number ofsub-frames per superframe are typically relatively low (e.g., usuallyless than 5).

To the extent that IBRs deployed in the field are within co-channelinterference range of each other and configured to use overlappingchannels in general or periodically, such IBRs advantageouslysynchronize their TDD fixed superframe timing to minimize simultaneousco-channel TxSF/RxSF overlaps amongst disparate links. Several exemplarysynchronization strategies may be used by such IBRs to align superframetiming boundaries in such scenarios. For example, if “free-running” or“self-synchronizing” IBRs are able to detect at least a preamble, atraining block, a PLCP header, an unencrypted MPDU header, or otherinformation such as management frames with timing information that maybe decipherable without link-specific encryption keys that correspond tolinks involving a different AE-IBR, then the slower in time IBRs mayadopt the superframe timing boundary and cadence of the faster in timeIBRs. At the AE-IBR, which may act as a local timing master, this can beperformed directly by making a timing offset and communicating it to isassociated RE-IBR(s). At the RE-IBR, which may be able to detectdisparate link information otherwise undetectable at its associatedAE-IBR, the RE-IBR can inform its AE-IBR via a management frame of anytiming offset necessary to obtain local disparate AE-IBR co-channelsuperframe timing synchronization. It will be appreciated that thisprocess is ongoing because after synchronizing, the reference clocks inthe AE-IBRs inevitably will drift differently over time.

In certain field deployment scenarios, IBRs located in the same regionalarea may be capable of undesirably interfering with each other at rangesbeyond their ability to detect and synchronize as described above. Analternative synchronization strategy better suited to this situationwould utilize a network-wide central synchronization capability. Oneexample of this would be the use of Global Positioning Satellite (GPS)timing at each AE-IBR. GPS is more than adequate in terms of timingaccuracy for the needs of synchronizing superframe timing boundaries.However, GPS adds cost, size, power consumption and form factorconstraints that may be undesirable or unacceptable to some IBRs. Notefurther that because IBRs are designed to operate in street leveldeployments with obstructed LOS, GPS may fail to operate in places whereRE-IBRs function normally. Another alternative would be to use asystem-wide synchronization technique such as SynchE or IEEE 1588v2. Inthis scenario, AE-IBRs are configured to derive timing parameters in aconsistent fashion. Alternatively, the AE-IBRS include IBMS Agentscapable of coordinating such configurations when co-channel operation ina mutual interfering deployment is encountered.

In the deployment scenario where multiple AE-IBRs are co-located (e.g.,at a single pole, tower, building side or rooftop), even if such IBRsare configured to avoid co-channel operation, at least some form oflocal superframe timing synchronization for TDD may be utilized to avoidoverloading receiver RF/analog circuits from simultaneous in-bandtransmissions and receptions amongst the co-located IBRs. One exemplarystrategy for the above synchronization would be to distribute by hardwiring a local superframe timing synchronization signal which can beconfigured at or arbitrarily assigned to any one of the co-located IBRs.

Note that the foregoing descriptions and figures for exemplary IBRembodiments have provided minimal internal details on the distributionof various clocks and timing references amongst the structural andfunctional elements of the IBR. These exemplary embodiments can all berealized using substantially conventional strategies for internal timingdistribution.

As described above, IBRs with fixed superframe timing may use a NACKprotocol wherein a previous superframe FCS failure in receive causesNACK=1 in an MPDU header of the respective link in transmitting its nextsequential superframe PPDU back to the sender of the MPDU received inerror. If the original sender detects NACK=1, then a re-transmission ofthe previous superframe PPDU contents occurs at the direction of its MACTx Buffer and Scheduler 3308. Otherwise, the original sender discardsthe previous superframe PPDU contents at its MAC Tx Buffer and Scheduler3308. This approach is different from many conventional wireless datanetworking protocols that use ACK receipt to guarantee 100% delivery atthe MAC layer even if theoretically unbounded re-try attempts may berequired. This fixed superframe NACK protocol is similar to using an ACKprotocol with a “time to live” of effectively one superframe durationafter initial transmission. This approach advantageously bounds thelatency at a very low length for processing frames through the IBR MACwithout resorting to reliability of simply the raw frame error rate. Byallowing one immediate re-transmission opportunity, this fixedsuperframe NACK protocol effectively produces a net frame error ratethat is the square of the raw frame error rate. For example, if the rawframe error rate were 10⁻³ (1 in 1000), the net frame error rate per thefixed superframe NACK protocol should be approximately 10⁻⁶ (1 in1,000,000).

To improve the reliability of the fixed superframe timing NACK protocoleven further, IBRs may have the TxPLCP set a 1 bit field in the PLCPheader(s) for the Modulator Core j corresponding to the LID with NACK=1(or with a previous RxSF(s) PLCP header FCS failure as described above).This approach advantageously exploits the fact that PLCP headers,typically sent at the most reliable MCS, are of short duration andalways sent immediately after the Training Block 0 (see FIG. 26) wherethe channel equalization coefficients most accurately reflect currentconditions. Thus, a NACK bit sent in a PLCP header is likely to bereceived accurately even if the PPDU payload that includes the MPDUheader NACK field is not. In the case where NACK=1 in a PLCP header, anexemplary PLCP Header Analyzer 3004 (see FIG. 30) will signal theexemplary MAC Tx Buffer and Scheduler 3308 via the IBR Control 3352. Inthe case where an originator of an MPDU does not receive a valid NACKfield in either the PLCP header or the next MPDU header of a subsequenttransmit superframe, then the MAC Tx Buffer and Scheduler 3308 maychoose to re-send the pending MPDU(s), possibly at a more reliable MCSas directed by the RLC 656, or may choose to discard the pending MPDU(s)so as not to cascade latency for further frames awaiting transmission.Such a decision may be made based on configurable policies that may varywith current conditions or be updated by the IBMS Agent 700. To theextent that a “blind” retransmission is made due to an invalid NACKfield wherein the actual NACK field value was NACK=0, then the receivepath MPDU Header Analyzer 3336 at the destination IBR will discard there-sent MPDU(s) based on duplicate detection of the FSN at the MPDUHeader Analyzer 3336 shown in FIG. 33. The MPDU Header Analyzer 3336 mayalso report incidents of duplicate detections of MPDUs for a given LIDto the RRC 660 and RLC 656 as shown in FIG. 33. This provides anadvantageous additional local indication to the RRC 660 and RLC 656 ofproblems being encountered in the transmit direction from such an IBRfor the particular LID.

With reference to FIGS. 31 and 32, the RRC 660 and RLC 656 interact withthe IBR MAC 612 and various elements of the IBR PHY both via “normal”frame transfers and direct control signals via the conceptual IBRControl plane. Both the RRC 660 and the RLC 656 may execute concurrentcontrol loops with the respective goals of optimizing radio resourceallocations and optimizing radio link parameters for current resourcesin view of the dynamic propagation environment conditions, IBR loading,and possibly system-wide performance goals (via the optional IBMS Agent700). It is instructive to view the RLC 656 as an “inner loop”optimizing performance to current policies and radio resourceallocations for each active link and to view the RRC 660 as an “outerloop” determining if different policies or radio resource allocationsare desirable to meet overall performance goals for all IBRs currentlyinteracting with each other (intentionally or otherwise). Typically boththe RRC 660 and the RLC 656 are implemented as software modulesexecuting on one or more processors.

The primary responsibility of the RLC 656 in exemplary IBRs is to set orcause to be set the current transmit MCS and output power for eachactive link. In one exemplary embodiment described above, the RLC 656provides information to the TxPLCP that enables, for example, a PLCPController 2812 in FIG. 28 to set the MCS for a particular LID. Inanother embodiment compatible with the single MPDU for fixed superframetiming with NACK protocol example of IBR MAC operation, the RLC 656determines the MCS for each LID and then communicates it directly to theMAC Tx Buffer and Scheduler 3308 of FIG. 33 and the PLCP Controller 2812for every TxPLCP Mod-j of FIGS. 27 and 28. The RLC 656 causes thetransmit power of the IBR to be controlled both in a relative senseamongst active links, particularly of interest for the AE-IBR in a PMPconfiguration, and also in an overall sense across all transmits chainsand antennas. In an exemplary IBR embodiment, the RLC 656 can cause suchtransmit power control (TPC) as described above by setting parameters atthe Channel Equalizer Coefficients Generator 2332 of FIG. 23 (forrelative power of different simultaneous modulation streams), at eachactive transmit RF chain Tx-m 636 of FIGS. 6, 7 and 16 (for relativepower of different simultaneous RF chains) and at each Front-end PA1104, 1204 within the IBR Antenna Array of FIGS. 6, 7, 10-12 (for totalpower from all antennas).

In some embodiments, the RLC 656 can determine its MCS and TPCselections across active links based on information from various sourceswithin the IBR. For example, the IBR MAC can deliver RLC control framesfrom other IBRs with information from such other IBRs (for example,RSSI, decoder metrics, FCS failure rates, etc.) that is useful insetting MCS and TPC at the transmitting IBR. Additionally, such RLCcontrol frames from an associated IBR may directly request or demandthat the RLC in the instant IBR change its MCS and/or TPC values fortransmit directly on either a relative or absolute basis. For TDD IBRdeployments, symmetry of the propagation environment (in the absence ofinterference from devices other than associated IBRs) makes receiverinformation useful not only for sending RLC control frames to thetransmitting IBR but also for use within the receiving IBR to set itstransmitter MCS and/or TPC. For example, the FCS Pass? analyzer 3344 andMPDU Header Analyzer 3336 of the exemplary IBR MAC in FIG. 33 can supplythe RLC 656 with useful information such as FCS failures, duplicatedetections and ACK or NACK field values. Similarly, each PLCP HeaderAnalyzer 3004 as shown in FIG. 30 can send the RLC FCS failure status onthe PLCP Header and/or PPDU payload(s). Another possibility is to havethe PLCP Header carry a closed-loop TPC control field requesting a TPCstep up or down or stay the same. Additionally, the Channel EqualizerCoefficients Generator 2332 in computing channel weights can provide SNRand/or SINR information per demodulator stream that can help the RLC setMCS and/or TPC as shown in FIG. 23. Each decoder within each demodulatorcore j can provide the RLC 656 with decoder metrics useful for MCSand/or TPC as indicated by FIGS. 20 and 22 for example. And each receivechain Rx-n 640 of FIG. 17 can provide receive signal strength indicator(RSSI) values helpful especially for determining TPC requests back tothe transmitting IBR.

The actual MCS values are typically selected from a finite length listof modulation types and coding rates. Exemplary IBRs can use QAM rangingfrom 2-QAM (better known as BPSK), through 4-QAM (better known as QPSK),16-QAM, 64-QAM, 256-QAM and 1024-QAM. Exemplary IBRs can use a basecoding rate of ⅓ or ½ and then can use “puncturing” (whereinpredetermined bit positions are deleted in transmit and replaced bydummy bits in receive) to derive a set of effective coding rates of, forexample only, ½, ⅔, ¾, ⅚, ⅞, and 9/10. In typical embodiments, thelowest MCS index corresponds to the lowest available QAM constellationsize and the lowest available coding rate (i.e. the most reliabletransmission mode) and the highest MCS index corresponds to theconverse.

The TPC absolute range tends to be lower for IBRs than that desired formany conventional wireless networking systems operating in obstructedLOS due to the more limited range of separations between AE-IBRs andRE-IBRs for backhaul applications (i.e. backhaul radios are almost neverplaced in close proximity to each other). The relative variation indesired power between active links at an AE-IBR may also limited inrange particularly by the transmit DACs.

Many possible algorithms are known for generally relating informationprovided to the RLC 656 as described above to selecting MCS and TPCvalues. In dynamic propagation environments, averaging or dampeningeffects between channel quality information and MCS changes areadvantageously utilized to avoid unnecessarily frequent shifts in MCS.To the extent that an IBR is operating at below the maximum allowableTPC value, it is generally advantageous to permit TPC to vary morequickly from superframe to superframe than the MCS. However, ifoperating at maximum allowable TPC, then it is often advisable toimmediately down select MCS to a more reliable setting upon detection ofan MPDU FCS failure and/or a NACK=1 condition. Conversely, up selectingMCS is usually performed only after repeated superframe metricsindicating a high likelihood of supporting an MCS index increase. At thelimit, where RLC 656 has reached maximum TPC and minimum MCS (mostreliable mode), to maintain ongoing link reliability, the imperativeincreases for the RRC 660 to allocate different resources to enable theRLC 656 to operate again in MCS and TPC ranges with margin for temporalchannel impairment.

The primary responsibility of the RRC 660 is to set or cause to be setat least the one or more active RF carrier frequencies, the one or moreactive channel bandwidths, the choice of transmit and receive channelequalization and multiplexing strategies, the configuration andassignment of one or more modulated streams amongst one of moremodulator cores, the number of active transmit and receive RF chains,and the selection of certain antenna elements and their mappings to thevarious RF chains. Optionally, the RRC may also set or cause to be setthe superframe timing, the cyclic prefix length, and/or the criteria bywhich blocks of Training Pilots are inserted. The RRC 660 allocatesportions of the IBR operational resources, including time multiplexingof currently selected resources, to the task of testing certain linksbetween an AE-IBR and one or more RE-IBRs. The RRC 660 evaluates suchtests by monitoring at least the same link quality metrics as used bythe RLC 656 as evident in the exemplary embodiments depicted in FIGS. 6,7, 17, 18, 20, 22, 23, 29, 30, 31, 32 and 33. Additionally, in someembodiments, additional RRC-specific link testing metrics such as thoseproduced by the exemplary Training Data Analyzer described above forFIG. 30 in relation to the exemplary Training Data Library of FIG. 28are used. The RRC 660 can also exchange control frames with a peer RRCat the other end of an instant link to, for example, provide certainlink testing metrics or request or direct the peer RRC to obtain linkspecific testing metrics at the other end of the instant link forcommunication back to RRC 660.

In some embodiments, the RRC 660 causes changes to current resourceassignments in response to tested alternatives based on policies thatare configured in the IBR and/or set by the optional IBMS Agent 700 asdepicted for example in FIGS. 7, 32, and 33. An exemplary policyincludes selecting resources based on link quality metrics predicted toallow the highest throughput MCS settings at lowest TPC value.Additional exemplary policies may factor in minimizing interference bythe instant link to other AE-IBR to RE-IBR links (or other radio channelusers such as conventional PTP radios) either detected at the instantIBRs or known to exist at certain physical locations nearby as set inconfiguration tables or communicated by the optional IBMS Agent 700.Such policies may also be weighted proportionately to reach a blendedoptimum choice amongst policy goals or ranked sequentially inimportance. Also as discussed above regarding the RLC 656, the RRC 660may have policies regarding the fraction of IBR resources to be used forRRC test purposes (as opposed to actual IBR backhaul operations) thatfactor the current RLC settings or trajectory of settings.

The RRC 660 in one IBR can communicate with the RRC in its counterpartIBR for a particular link AE-IBR to RE-IBR combination. For example, theRRC 660 sends RRC control frames as discussed for the exemplary IBR MACof FIG. 33 or invokes a training mode within the exemplary Tx PLCP asdiscussed for FIG. 28.

In some embodiments, for either PTP or PMP deployment configurations,the selection of either the one or more active RF carrier frequenciesused by the RF chains of the IBR RF, the one or more active channelbandwidths used by the IBR MAC, IBR Modem, IBR Channel MUX and IBR RF,the superframe timing, the cyclic prefix length, or the insertion policyfor blocks of Training Pilots is determined at the AE-IBR for any givenlink. The RE-IBR in such an arrangement can request, for example, an RFcarrier frequency or channel bandwidth change by the AE-IBR by sendingan RRC control frame in response to current link conditions at theRE-IBR and its current RRC policies. Whether in response to such arequest from the RE-IBR or due to its own view of current linkconditions and its own RRC policies, an AE-IBR sends the affectedRE-IBRs an RRC control frame specifying at least the parameters for thenew RF frequency and/or channel bandwidth of the affected links as wellas a proposed time, such as a certain superframe sequence index, atwhich the change-over will occur (or alternatively, denies the request).The AE-IBR then makes the specified change after receiving confirmationRRC control frames from the affected RE-IBRs or sends a cancellation RRCcontrol frame if such confirmations are not received before thescheduled change. In some deployment situations, the RRC policycondition causing the change in the RF carrier frequency and/or channelbandwidth for a particular LID may be a directive from the IBMS Agent700.

The selection of other enumerated resources listed above at an IBR cangenerally be made at any time by any given IBR of a link. Additionally,requests from the opposite IBR can also be made at any time via RRCcontrol frames. An RE-IBR typically attempts to utilize all availablemodulator and demodulator cores and streams as well as all available RFchains to maximize the robustness of its link to a particular AE-IBR. Inan RE-IBR embodiment where at least some redundancy in antenna elementsamongst space, directionality, orientation, polarization and/or RF chainmapping is desirable, the primary local RRC decision is then to chooseamongst these various antenna options. An exemplary strategy forselecting antenna options (or other enumerated resources listedpreviously) at the RE-IBR is to apply alternative selections of suchresources to the Training Data portion of PPDUs described above inrelation to the Tx PLCP and FIGS. 27 and 28. The RRC 660 at the RE-IBRthen compares link quality metrics such as those used by the RLC 656 andtraining data metrics from the exemplary Training Data Analyzer of FIG.30 to determine if the alternatively-selected resources are likely toresult in improved performance based on current IBR RRC policiescompared to the known performance of the currently-selected resources ofthe instant link. If the RRC 660, based on one or more such alternativeselection tests, decides that the instant link's performance goals arelikely to improve by using the alternately-selected resources, then theRRC 660 at such RE-IBR can cause such selections to become the newcurrent settings at any time and without requiring notification to theAE-IBR.

For the RE-IBR alternate resource selection process described aboveapplied to a TDD configuration, channel propagation symmetry for a givenlink (if interference is ignored) may make changing to a correspondingset of resources for transmit from such RE-IBR as have beenalternatively-selected for receive preferable. However, this isgenerally not true for an FDD configuration or a scenario where unknowninterference represents a significant channel impairment or where a PMPAE-IBR has simultaneous links to other RE-IBRs. In such scenarios, anRE-IBR may notify the AE-IBR when such RE-IBR is testingalternately-selected resources in a portion of its Tx PPDUs, whether inresponse to an RRC control frame request by the AE-IBR or by suchRE-IBR's own initiative, and then receive a return RRC control framefrom the AE-IBR that either reports the measured link quality metricsobserved at the AE-IBR and/or directs the RE-IBR to adopt thealternatively-selected resources for future Tx PPDUs from the RE-IBR onsuch link.

For the PTP configuration, an AE-IBR performs its RRC-directed alternateresource selections using substantially the same processes describedabove for the RE-IBR but with the roles reversed appropriately. In thePMP configuration, an AE-IBR may utilize similar RRC-directed testing ofalternate resource selections across its multiple current links but tothe extent that such links depend concurrently on certain resources, thedecision to actually change to different resources may be based onpolicies applied to the benefit of all current links. One strategy forPMP operation of IBRs is to use the maximum possible RF chain andantenna element resources at all times at the AE-IBR and then optimizeselectable resources at the RE-IBRs to best achieve RRC policy goals.

Note that in some deployment situations, spectrum regulations, such asthose set by the Federal Communications Commission (FCC) in the USA, mayrequire active detection of and avoidance of interference with otherusers of the spectrum (such as, for example, radar systems). The processof detecting such other co-channel spectrum users and then changing RFcarrier frequencies to another channel void of such other uses iscommonly called Dynamic Frequency Selection (DFS). Spectrum regulationsmay require that a DFS capability operate in a certain manner inresponse to certain interference “signatures” that may be detected atthe receiver of a certified radio for such spectrum. For example, someregulations require that upon detection of certain pulse lengths andreceived powers that certified radios change immediately to anotherchannel known from some minimum observation time not to have suchinterferers in operation. In some exemplary IBR implementations, suchobservations of alternative channels in advance of needing to make achange can be performed by time division multiplexing certain RF chainand antenna resources to make such measurements using RSSI and/orchannel equalization metrics reported to the RRC 660. In someembodiments, the AE-IBR and the one of more associated RE-IBRscoordinate such observations using RRC control frames to minimallydisrupt backhaul operations and maximally increase the aggregateobservation time and improve the observation accuracy. In otherexemplary IBR embodiments, at least one IBR, typically the AE-IBR, hasat least one Rx-n chain, one antenna and possibly one demodulator corepath through the IBR Channel MUX dedicated to such spectralobservations.

In embodiments with the optional IBMS Agent 700, the above channelobservation techniques of the RRC 660 can also be used in a “probe inspace” mode of operation, either at one IBR or coordinated amongstmultiple IBRs, to observe and record RF channel activity withindesignated portions of the addressable bands of operation. Such spectralanalysis information may be passed from the RRC 660 to the IBMS Agent700 for further analysis and possibly communication to other IBMS Agentsor to other databases or expert systems elsewhere within the IBRoperator's private network or to an external network such as theInternet.

Note also that the DFS operation described above is desirable forexemplary IBRs operating in spectrum bands that do not require DFSexplicitly. For example, if IBRs are deployed in licensed spectrum whereconventional PTP links operate, such conventional links generally lackthe RF carrier frequency agility and radio resource control intelligenceof the IBR. Even if the interference immunity capabilities of the IBRthrough the advantageous combinations of elements described herein issufficient to reject the interference caused by such conventional PTPlinks at the IBR receiver, it is still desirable to have the RRC 660perform DFS to avoid the converse scenario where the IBRs areinterfering with the conventional PTP links. This may be advantageousbecause it minimizes licensed band user conflicts especially if adifferent operator uses the conventional PTP equipment from thatoperating the IBRs. The presence of the conventional PTP link may bedetected in normal operation of the one or more IBRs using a particularRF carrier frequency channel or may be communicated to the RRC 660 viathe optional IBMS Agent 700 that has gathered the information fromanother source. An exemplary technique that the RRC 660 can use in anIBR where N>L and the instant SINR is approximately the same as the SNR(i.e. no significant co-channel interference) per metrics available tothe RRC 660 from the Channel Equalizer Coefficients Generator 2332 is toassign up to N minus L combinations of an antenna element 652, an Rx-nchain, and a channel MUX receive path to a Complex DFT-n to differentfrequency channels than the instant link channel to perform DFS or“probe in space” measurements and spectral analysis. For FDDconfigurations, assigning these combinations to monitor the instant orcandidate transmit frequency channels (possibly during a time when thetransmitter is otherwise inhibited) can allow the RRC to evaluatepotential interference to other conventional PTP links and to adjusttransmit resources accordingly. To the extent that the remaining atleast L receive chains provide sufficient SNR or SINR to maintain theinstant traffic load, this approach allows the RRC 660 to utilizeavailable IBR resources simultaneously for both supporting link trafficand supporting DFS or “probe in space” measurements and spectralanalysis.

As described previously, exemplary IBRs advantageously exploit thepropagation path diversity usually present in an obstructed LOSenvironment to send multiple modulated streams concurrently and thusincrease overall link throughput. For practical reasons regarding actualfield deployments, it is likely that some IBRs will be deployed inlocations where propagation may be dominated at least at some times byunobstructed LOS conditions. In such situations, IBR embodiments usingthe IBR Channel MUX 628 of FIG. 23 may not be able to resolve multiplestreams during the training Block 0 of FIG. 26 because all streamsarrive at all antennas by substantially similar paths which makescancellation for multiple demodulation streams impractical. Onealternative for the RRC 660 in this situation is to allocate one streamper link only. However, this results in reduced throughput for the linkwhich is not only undesirable but highly counterintuitive for manybackhaul operations personnel likely to perceive that unobstructed LOSshould be “better” than obstructed LOS. Thus, the RRC 660 may alsoevaluate certain antenna selection options to intentionally create pathdiversity in otherwise unobstructed or nearly unobstructed LOSconditions.

A first alternative for the RRC 660 to provide multiple streams withboth obstructed and non-obstructed LOS operation is the dynamic testingand possibly selection of mapping different modulator streams todifferent antenna elements (via separate RF chains) based on differentantenna polarizations. Because of the typically substantial signalimpairment associated with a link that is transmitting receiving fromopposite polarization antenna elements, testing of alternativepolarization antenna elements with training data may be pre-arranged intime by RRC control frames exchanged by both IBRs with an instant link.Similarly, the AE-IBR may select any changes in link antenna elementsinvolving polarization and verify an agreed upon changeover time by RRCcontrol frame exchange for reasons analogous to those for RF carrierfrequency or channel bandwidth changes. A significant advantage of usingpolarization diversity amongst the set of selectable antenna elements isthat multiple stream throughput can be maintained using a common set ofchannel equalization techniques as described above for MIMO operationwith the exemplary IBR Channel MUX of FIG. 23.

A second alternative for the RRC 660 to provide multiple streams withboth obstructed and non-obstructed LSO operation is the dynamic testingand possibly selection of mapping different modulator streams todifferent antenna elements (via separate RF chains) based on differentdirection orientations of antenna elements as is possible with exemplaryantenna arrays such as those depicted in FIGS. 14 and 15. The RRC 660 insuch IBR embodiments can test and/or change-over such multi-directionalantenna element combinations entirely at one end of a link without thenecessity of exchanging RRC control frames or requiring a coordinatedchange-over at both IBRs in a link simultaneously. For at least thepreceding reason, this second alternative to maintaining multi-streamoperation by using multi-directional antenna element combinations thatintentionally create propagation path diversity may be desirable for anRE-IBR used in a PMP configuration. An example antenna array suitablefor such an RE-IBR is depicted in FIG. 14. Furthermore, this directionalorientation diversity strategy to multi-stream operation in otherwiseunobstructed LOS is also advantageous compared to the polarizationdiversity option in PMP deployments because it does not require that theAE-IBR deploy certain resources (e.g., dedicated RF chains and antennaelements) to link modalities of benefit to some RE-IBRs but not others(that do experience obstructed LOS) as may occur with polarizationdiversity in a PMP deployment.

With reference to FIGS. 7, 32 and 33, the Intelligent BackhaulManagement System (IBMS) Agent 700 is an optional element of the IBRthat optimizes performance of the instant links at the IBR as well aspotentially other IBR links in the nearby geographic proximity includingpotential future links for IBRs yet to be deployed. As described abovein reference to the RRC 660 and depicted in FIG. 32, the primaryinteraction of the IBMS Agent 700 with the internal elements of the IBRis via a direct bi-directional path to the RRC 660. In one direction,the various policies and configuration parameters used by the RRC 660 toallocate resources within and amongst IBRs with active links to eachother are sent from the IBMS Agent 700 to the RRC 660. In the returndirection, the RRC 660 reports operational statistics and parametersback to the IBMS Agent 700 both from normal operation modes and from“probe in space” modes as directed by the IBMS Agent 700.

In contrast with the RRC 660, which communicates with other elements ofthe IBR internally or with other RRC entities at IBRs actively linked toits IBR, the IBMS Agent 700 can receive information from or transmitinformation to or initiate sessions with other elements of the overallIBMS that are logically located anywhere within any network (subject toappropriate access privileges). As shown in FIG. 32, the IBMS Agent 700appears to the overall network as an Applications layer entity thatexchanges messages typically over TCP/IP and any of the link layerinterfaces within the IBR. For the common deployment of IBRs within acellular system Radio Access Network (RAN) to backhaul cellular basestation sites, it may be necessary for the IBR to tunnel TCP/IP across acellular RAN specific transport and network layers protocol, such asGPRS Tunneling Protocol (GTP) to reach a gateway that bridges to TCP/IPand on to the desired other IBMS elements.

In some embodiments, the IBMS Agent 700 can act as an autonomous entitythat per configuration settings draws information from network resources(whether private or public) and serves as an expert system local to be aspecific IBR to optimize its performance. In other embodiments, the IBMSAgent 700 interacts with other peer IBMS Agents at other IBRs within its“interference” zone of influence via self-discovery within the immediatenetwork so that such peer IBMS Agents can collectively optimize IBRperformance within the zone. In some embodiments, the IBMS Agent 700 isa client of one or more IBMS servers that may be within the privateand/or public networks such that communications with a particular IBMSAgent 700 is always to or from or via such IBMS servers.

The information gathered at the IBR and distilled by the IBMS Agent 700regarding, for example, operational statistics (such as RSSI, channelequalization metrics, FCS failures, etc.) and resource selections (suchas antennas, channel bandwidth, modulator stream assignments, etc.), maybe sent to an IBMS server. Such information can be used by the IBMSserver to improve performance predictability of future IBR deploymentsor to enable overall IBR system performance of all links within thepurview of the IBMS server by policy optimization across IBRs. Thecommunications from the IBMS server to the IBMS Agents can include suchoptimized policy parameters based on information from other IBMS Agentsand private and/or public databases such as directories of known non-IBRlinks including their locations, antenna patterns, power levels, carrierfrequencies and channel bandwidths, tower heights, etc.

With reference to FIGS. 31 and 32, other exemplary applications layerprotocols are shown for the IBR. A Hyper Test Transfer Protocol (HTTP)server application can enable any browser (or browser-like application)on a networked computer, including a WiFi connected mobile device suchas laptop, table or smart-phone, to query or modify various IBRconfiguration parameters. The Simple Network Management System client ofFIGS. 31 and 32 is an example of an industry-standard network managementutility that can also be used to query or modify IBR configurationparameters using tools such as HP Open View.

The foregoing description of the various elements of the IBR inreference to FIGS. 5-36 have described numerous exemplary embodiments.These exemplary element embodiments may be assembled together in manydifferent combinations and permutations. A very short description of buta few of the possible overall IBR exemplary embodiments is summarizedbriefly below.

A first exemplary embodiment of an IBR includes the features outlined inTable 1.

TABLE 1 First Exemplary Intelligent Backhaul Radio Features TDDoperation PTP configuration only (AE-IBR or RE-IBR) SC-FDE modulationJmod = Jdem = 1 K =2, M = 4 L = 2, N = 4 Q = 8, antenna array similar toFIG. 14 front facet with 2 Vertical and 2 Horizontal polarizationantenna elements at 15 dBi side facets each with 1 Vertical and 1Horizontal polarization antenna elements at 12 dBi QPSK, 16 QAM, 64 QAM,256 QAM ⅓, ½, ⅔, ¾, ⅚, ⅞ coding rates 1 or 2 modulated streams MMSEcombining of up to 4 receive chains MRC weighting of up to 4 transmitchains based on blended weights per chain across entire channel asderived from Rx conditions fixed superframe (1 ms) NACK protocol up to28 MHz channel bandwidth single MPDU per PPDU

This first exemplary embodiment can have a very high MAC efficiency(ratio of MPDU payload bits to overall MAC bits) under heavy loadingfrom the IBR LLC—in excess of 95%. Furthermore, the PHY efficiency(ratio of time where PPDU payload symbols excluding PAD are actuallytransmitted to superframe time) can exceed 90% for typical channelimpairment conditions and ranges of 2 km or less. At 28 MHz symbol rateand channel bandwidth, 256 QAM, 2 modulated streams, ⅞ rate coding, andwith MAC and PHY efficiencies of 95% and 90% respectively, the aggregatebi-directional throughput for this first exemplary IBR embodiment canexceed 330 Mb/s with an average end to end latency of about 2 ms.

A second exemplary embodiment of an IBR includes the features outlinedin Table 2.

TABLE 2 Second Exemplary Intelligent Backhaul Radio Features TDDoperation PMP configuration as the AE-IBR OFDM modulation Jmod = 2, Jdem= 3 K =2, M = 4 L = 5, N = 5 Q = 9, antenna array similar to a 180°azimuth coverage “half-array” of FIG. 15 per each of 4 facets: 1Vertical and 1 Horizontal polarization antenna elements at 15 dBi on“top”: 1 non-polarized, 180° azimuth (aligned with array) coverage 8 dBielement connected only to the 5^(th) Rx chain and 3^(rd) demodulatorcore QPSK, 16 QAM, 64 QAM, 256 QAM ⅓, ½, ⅔, ¾, ⅚, ⅞ coding rates up to 2modulated streams per modulator core MMSE combining of up to 5 receivechains Eigenbeamforming transmit SDMA of up to 4 transmit chains fixedsuperframe (0.5 ms or 1 ms) NACK protocol TDMA alternate superframe perRE-IBR up to 2 RE-IBRs per modulator core up to 28 MHz channel bandwidthsingle MPDU per PPDU

This second exemplary embodiment uses OFDM rather than SC-FDE in orderto enable transmit SDMA of up to 4 RE-MRs (only 2 simultaneously) in ahighly frequency selective channel. As discussed above, SC-FDE couldalso be used in this PMP AE-IBR with theoretically similar performancebut with more complex baseband processing required. This secondexemplary embodiment should have similar MAC efficiency to the first forthe 1 ms superframe case but the PHY efficiency (which needs adefinition that accounts for data block pilot subchannels andzero-padded subchannels) is typically lower with 85% being excellent.The eigenbeamforming may also require additional overheads in manypropagation environments. If 4 RE-MRs are used in TDMA mode, the latencywould expand to 4-5 ms on average for 1 ms superframes and about halfthat for 0.5 ms superframes. With 2 RE-IBRs that are spatially separableand with operating parameters set as described for the first exemplaryembodiment above, aggregate bi-directional throughput for this secondexemplary embodiment can be as high as about 600 Mb/s. Note that the“top” antenna which is not connected to a transmit chain can be used toprovide the MMSE combiner with an additional degree of freedom to cancelinterference when 4 modulated streams from 2 RE-MRs are beingsimultaneously received. It can also be used advantageously as a “probein space” to provide information to the IBMS or to assist in DFS byscanning channels not currently used at the AE-IBR. Also note thatalthough this second exemplary embodiment can be used as an RE-IBR foritself that preferably this role may be filled by the third exemplaryembodiment described below.

A third exemplary embodiment of an IBR includes the features outlined inTable 3.

TABLE 3 Third Exemplary Intelligent Backhaul Radio Features TDDoperation PTP configuration only (AE-IBR or RE-IBR) or PMP configurationas the RE-IBR OFDM modulation Jmod = Jdem = 1 K = 2, M = 4 L = 2, N = 4Q = 8, antenna array similar to FIG. 14 front facet with 2 Vertical and2 Horizontal polarization antenna elements at 15 dBi side facets eachwith 1 Vertical and 1 Horizontal polarization antenna elements at 12 dBiQPSK, 16 QAM, 64 QAM, 256 QAM ⅓, ½, ⅔, ¾, ⅚, ⅞ coding rates 1 or 2modulated streams MMSE combining of up to 4 receive chains MRC weightingof up to 4 transmit chains based on blended weights per chain acrossentire channel as derived from Rx conditions fixed superframe (1 ms)NACK protocol up to 28 MHz channel bandwidth single MPDU per PPDU

Note that while this third embodiment can also be used as a PTP AE-IBRand RE-IBR combination, it is unlikely to provide meaningful performanceimprovements for PTP compared to the first exemplary embodiment andquite possibly would have slightly lower aggregate bi-directionalthroughput and would require a less efficient and more expensive poweramplifier and stricter phase noise considerations. Forcommercially-available components today, using OFDM versus SC-FDE at RFcarrier frequencies above 10 GHz is extremely challenging. Note that forbelow 10 GHz operation, it is commercially feasible today to use SDRbaseband approaches and common chain and Front-end components to buildan IBR software programmable as either the PTP first exemplaryembodiment or the PMP RE-IBR third exemplary embodiment.

A fourth exemplary embodiment of an IBR includes the features outlinedin Table 4.

TABLE 4 Fourth Exemplary Intelligent Backhaul Radio Features FDDoperation PMP configuration as the AE-IBR OFDM modulation Jmod = 4, Jdem= 6 K = 8, M = 8 L = 10, N = 10 Q = 18, antenna array similar to FIG. 15per each of 8 facets: 1 Vertical and 1 Horizontal polarization antennaelements at 15 dBi on “top”: 2 non-polarized, opposite facing, 180°azimuth coverage 8 dBi elements connected only to the respective ones ofthe 9^(th) and 10^(th) Rx chains and the 5^(th) and 6^(th) demodulatorcores QPSK, 16 QAM, 64 QAM, 256 QAM ⅓, ½, ⅔, ¾, ⅚, ⅞ coding rates up to2 modulated streams per modulator core MMSE combining of up to 10receive chains Closed-loop eigenbeamforming transmit SDMA of up to 8transmit chains fixed superframe (.5 ms or 1 ms) NACK protocol TDMAround-robin superframe order per RE-IBR up to 4 RE-IBRs per modulatorcore up to 28 MHz channel bandwidth single MPDU per PPDU

This fourth exemplary embodiment is similar to the second exemplaryembodiment except that it utilizes a larger 360° azimuth antenna arrayand FDD operation as well as 4 time slot per modulator TDMA to supportup to 16 RE-MRs with an aggregate bi-directional throughput of about 2Gb/s after the increased overhead in efficiencies of the system areaccounted for. Latency also increases proportionately if 4 TDMA slotsare used.

A fifth exemplary embodiment of an IBR includes the features outlined inTable 5.

TABLE 5 Fifth Exemplary Intelligent Backhaul Radio Features FDDoperation PMP configuration as the RE-IBR OFDM modulation Jmod = Jdem =1 K = 2, M = 4 L = 2, N = 4 Q = 8, antenna array similar to FIG. 14front facet with 2 Vertical and 2 Horizontal polarization antennaelements at 15 dBi side facets each with 1 Vertical and 1 Horizontalpolarization antenna elements at 12 dBi QPSK, 16 QAM, 64 QAM, 256 QAM ⅓,½, ⅔, ¾, ⅚, ⅞ coding rates 1 or 2 modulated streams MMSE combining of upto 4 receive chains Selected unweighted Tx antennas as directed byreceiver at opposite end fixed superframe (1 ms) NACK protocol up to 28MHz channel bandwidth single MPDU per PPDUThe primary application of this fifth exemplary embodiment is to serveas an RE-IBR for the fourth exemplary embodiment AE-IBR.

A sixth exemplary embodiment of an IBR includes the features outlined inTable 6.

TABLE 6 Sixth Exemplary Intelligent Backhaul Radio Features FDDoperation PTP configuration only (AE-IBR or RE-IBR) SC-FDE modulationJmod = Jdem = 1 K = M = 2 L = 2, N = 4 Q = 6, antenna array similar toFIG. 13 3 Vertical polarization antenna elements, 2 at 13 dBi and 1 at18 dBi, 3 Horizontal polarization antenna elements, 2 at 13 dBi, 1 at 18dBi QPSK, 16 QAM, 67 QAM, 256 QAM, 1024 QAM ⅓, ½, ⅔, ¾, ⅚, ⅞ 9/10 codingrates 1 or 2 modulated streams MMSE combining of up to 4 receive chainsselection of 2 antennas in transmit fixed superframe (1 ms) NACKprotocol up to 56 MHz channel bandwidth single MPDU per PPDU

This sixth exemplary embodiment provides high aggregate bi-directionalthroughput of up to about 1.8 Gb/s with moderate complexity relative toother IBRs. This sixth exemplary embodiment performs optimally inpropagation channels with only moderate obstructions compared tounobstructed LOS. FDD also provides <1 ms average latency.

A seventh exemplary embodiment of an IBR includes the features outlinedin Table 7.

TABLE 7 Seventh Exemplary Intelligent Backhaul Radio Features FDD/TDDhybrid operation PTP configuration only (AE-IBR or RE-IBR) SC-FDEmodulation Jmod = Jdem = 1 K = 4, M = 8 L = 4, N = 8 Q = 12, antennaarray similar to FIG. 14 per each of 3 facets: 2 Vertical and 2Horizontal antenna elements of 15 dBi each QPSK, 16 QAM, 64 QAM, 256 QAM⅓, ½, ⅔, ¾, ⅚, ⅞ coding rates up to 4 modulated streams MMSE combiningof up to 8 receive chains MRC weighting of up to 8 transmit chains basedon blended weights per chain across entire channel as derived from Rxconditions fixed superframe (1 ms) NACK protocol up to 56 MHz channelbandwidth single MPDU per PPDU

This seventh exemplary embodiment has additional resources compared tothe sixth exemplary embodiment to provide higher aggregatebi-directional throughput of about 3 Gb/s for a PTP link with <1 mslatency that can operate in a severely obstructed LOS propagationchannel. It advantageously uses hybrid FDD/TDD operation wherein eachfrequency duplexed channel alternates in opposite synchronization toeach other between transmit and receive. This enables a relativelystraightforward and efficient transmit chain weighting to be derivedfrom receive chain equalization analysis without increasing latency.Furthermore, an additional degree of frequency diversity (and space tothe extent different antennas are selected) is achieved. The FDD/TDDhybrid can be utilized on any FDD IBR deployment where spectrumregulations permit it. To the extent each FDD band operation relies onband-specific band-select filters in the Front-ends, then additionalcircuit complexity for switching between transmit and receive is needed.

Note that the preceding embodiments are a small subset of the possibleIBR embodiments that are enabled by the disclosure herein. Note furtherthat many additional optional structures or methods described herein canbe substituted within the above exemplary embodiments or otherembodiments.

For example, TDD CSMA could be used advantageously in high interferencespectrum allocations or where required by spectrum regulations as asubstitute for fixed superframe timing in the above exemplaryembodiments.

Note also that all of the above exemplary embodiments are compatiblewith any RF carrier frequencies in the range of interest fromapproximately 500 MHz to 100 GHz.

Note further that in multi-channel embodiments, it is possible to usedifferent access and MAC protocols in different channels especiallywhere advantageous for or required by spectrum regulations. For example,an IBR link may advantageously provide a “base” throughput capability ina channel expected to have minimal interference but for which licensingcosts or regulatory restrictions limit total throughput. Then a “surge”throughput capability can be provided in a second channel, such asunlicensed spectrum, where throughput can be higher but the risk oftemporal interference outages is also higher.

As evident from the above exemplary embodiments, OFDM is typically usedfor PMP deployments because at a baseband processing level it isrelatively less complex than SC-FDE if frequency-selective channels areto be used with transmit SDMA (at least at the AE-IBR). However, OFDMhas higher peak to average ratio and is more sensitive to carrierfrequency offset and phase noise than SC-FDE. Thus, for PTP, SC-FDE isoften preferable, especially for operation at RF carrier frequenciesabove 10 GHz where commercially viable components are expensive foreither OFDM power amplification or OFDM-compatible local oscillatorspecifications.

Note that in backhaul applications where links are nominally continuous,additional techniques can improve PHY efficiency. For example, withOFDM, the training block can be combined with a PLCP Header block byinterleaving subchannels appropriately. This is also possible for SC-FDEif DFT pre-coding (i.e. Tx block Assembler-k includes a DFT and Tx-Mux-mincludes an IDFT) is used. DFT pre-coding for SC-FDE can also be used topulse shape the transmitted waveform similar to OFDM zero padding and/orwindowing. The training block in SC-FDE can also be shorter than thedata blocks to save PHY overhead by either using another FFT fortraining or switching the FFT bin size during the training block. Thechannel equalization function so derived is then interpolated for use onthe longer data blocks with additional frequency bins. Also, when usingDFT pre-coding in SC-FDE, it is possible to time multiplex an FFT blockbetween transmit and receive or within transmit or receive such that allfour FFT operations of an SC-FDE transceiver can be realized by 2 oreven 1 FFT hardware core. Another technique to simplify PMP deploymentof IBRs is to use OFDM in the forward link from AE-IBR to the multitudeof RE-IBRs, and then use SC-FDE in the reverse link from RE-IBR toAE-IBR. This enables the advantages of transmit eigenbeamforming at theAE-IBR in a frequency-selective channel based primarily on receiveequalization while keeping the RE-IBRs relatively straightforward withmuch simpler transmitters than in the OFDM only case.

Numerous additional variations of the above-described elements of theIBR can also be advantageously utilized in substitution for or incombination with the exemplary embodiments described above. For example,antenna elements need not always be shared between transmit and receivewhether in TDD or FDD mode. In certain embodiments, it is preferable tohave a smaller number of transmit antenna elements, often with broaderazimuthal coverage than those used by the receive antenna elements, thatare always used versus a selectable larger number of receive antennaelements. In some embodiments with separate transmit and receive antennaelements, the respective front-ends of FIGS. 11 and 12 would not requirerespectively the SPDT switch or the band selection within a duplexerfilter and either the receive path elements for a front-end coupled to atransmit antenna element or the transmit path elements for a front-endcoupled to a receive antenna element, which further advantageouslyreduces cost and complexity of the IBR. In such embodiments, eachtransmit antenna element is coupled to a respective transmit poweramplifier which is in turn coupled to a respective transmit RF chainwherein such couplings may be selectable RF connections or fixed RFconnections—the latter for IBR embodiments wherein certain transmit RFchains are always connected to certain transmit power amplifiers andcertain transmit antenna elements whenever the IBR is in a transmitmode. Similarly, each receive antenna element is coupled to a respectivereceive low noise amplifier which is in turn coupled to a respectivereceive RF chain wherein such couplings are typically selectable RFconnections as described herein.

As another example, the NACK protocol described above with reference toFIG. 33 and various IBR embodiments can be extended to either ACK/NACK,single NACK as described above, or persistent NACK until delivered.Furthermore, such exemplary choices can be applied to individual MSDUsbased on either an indicator from the IBR LLC of FIG. 33 or inspectionof MSDU class of service or type of service header field bits as definedby a policy either within the IBR MAC or updateable via the optionalIBMS Agent 700. For example, MSDUs corresponding to Voice over InternetProtocol (VoIP) packets are typically sent at a high class of servicepriority but with so little tolerance for latency that it may bepreferable to send such MSDUs with no ACK/NACK retransmission option orwith only the single NACK protocol described previously. At the oppositeextreme, certain data transfers, such as for example only, cellularnetwork control or management plane messages or user file transfer data,may tolerate considerable or unpredictable latencies associated withpersistent retransmission until NACK=0 rather than rely on much slowerupper layer retransmission protocols. The policy for mapping MSDUs to aparticular ACK/NACK strategy may also be responsive to radio channelconditions and/or loading as determined within the various elements ofthe IBR described above. For example, when the current packet failurerate is very low and/or the loading demand on the IBR is high comparedto the capacity of the MCS, then one policy may be to minimizeretransmissions by transmitting most MSDUs with no ACK/NACK or a singleNACK. Alternatively, for a high packet failure situation and/or lowdemand at a given MCS, the opposite strategy may be used. Also, any ofthe IBR embodiments described herein that use copper Ethernet interfacesmay also use such interfaces to supply Power over Ethernet (PoE) to theIBR.

One or more of the methodologies or functions described herein may beembodied in a computer-readable medium on which is stored one or moresets of instructions (e.g., software). The software may reside,completely or at least partially, within memory and/or within aprocessor during execution thereof. The software may further betransmitted or received over a network.

The term “computer-readable medium” should be taken to include a singlemedium or multiple media that store the one or more sets ofinstructions. The term “computer-readable medium” shall also be taken toinclude any medium that is capable of storing, encoding or carrying aset of instructions for execution by a machine and that cause a machineto perform any one or more of the methodologies of the presentinvention. The term “computer-readable medium” shall accordingly betaken to include, but not be limited to, solid-state memories, andoptical and magnetic media.

Embodiments of the invention have been described through functionalmodules at times, which are defined by executable instructions recordedon computer readable media which cause a computer, microprocessors orchipsets to perform method steps when executed. The modules have beensegregated by function for the sake of clarity. However, it should beunderstood that the modules need not correspond to discreet blocks ofcode and the described functions can be carried out by the execution ofvarious code portions stored on various media and executed at varioustimes.

It should be understood that processes and techniques described hereinare not inherently related to any particular apparatus and may beimplemented by any suitable combination of components. Further, varioustypes of general purpose devices may be used in accordance with theteachings described herein. It may also prove advantageous to constructspecialized apparatus to perform the method steps described herein. Theinvention has been described in relation to particular examples, whichare intended in all respects to be illustrative rather than restrictive.Those skilled in the art will appreciate that many differentcombinations of hardware, software, and firmware will be suitable forpracticing the present invention. Various aspects and/or components ofthe described embodiments may be used singly or in any combination. Itis intended that the specification and examples be considered asexemplary only, with a true scope and spirit of the invention beingindicated by the claims.

The invention claimed is:
 1. A fixed wireless access radio forexchanging one or more data interface streams with one or more otherfixed wireless access radios, said radio comprising: a plurality ofreceive radio frequency (RF) chains, wherein at least a first subset ofthe plurality of receive RF chains is configured to convert from atleast a respective one of a plurality of receive RF signals within atleast a first receive frequency band to a respective one of a firstplurality of receive chain output signals, and wherein at least a secondsubset of the plurality of receive RF chains is configured to convertfrom at least a respective one of a plurality of receive RF signalswithin at least a second receive frequency band to a respective one of asecond plurality of receive chain output signals; a plurality oftransmit radio frequency (RF) chains, wherein at least a first subset ofthe plurality of transmit RF chains is configured to convert from atleast a respective one of a first plurality of transmit chain inputsignals to a respective one of a plurality of transmit RF signals withina first transmit frequency band, and wherein at least a second subset ofthe plurality of transmit RF chains is configured to convert from atleast a respective one of a second plurality of transmit chain inputsignals to a respective one of a plurality of transmit RF signals withina second transmit frequency band; and a plurality of antenna elements,wherein at least a first subset of the plurality of antenna elements isconfigured to operate over at least both of the first transmit frequencyband and the first receive frequency band and each antenna element ofthe first subset of the plurality of antenna elements is coupled orcouplable to at least one of the first subset of the plurality ofreceive RF chains or coupled or couplable to at least one of the firstsubset of the plurality of transmit RF chains, and wherein at least asecond subset of the plurality of antenna elements is configured tooperate over at least both of the second transmit frequency band and thesecond receive frequency band and each antenna element of the secondsubset of the plurality of antenna elements is coupled or couplable toat least one of the second subset of the plurality of receive RF chainsor coupled or couplable to at least one of the second subset of theplurality of transmit RF chains; wherein the fixed wireless access radiois configured to provide a base throughput capability using the firstreceive frequency band and the first transmit frequency band; andwherein the fixed wireless access radio is further configured to providea surge throughput capability using the second receive frequency bandand the second transmit frequency band.
 2. The fixed wireless accessradio of claim 1, wherein the first transmit frequency band iscoincident with the first receive frequency band.
 3. The fixed wirelessaccess radio of claim 1, wherein a first frequency band comprises atleast the first transmit frequency band and the first receive frequencyband.
 4. The fixed wireless access radio of claim 3, wherein the firstfrequency band is either within a frequency range of between 2 GHz and 7GHz or within a frequency range of above 10 GHz.
 5. The fixed wirelessaccess radio of claim 1, wherein the second transmit frequency band iscoincident with the second receive frequency band.
 6. The fixed wirelessaccess radio of claim 1, wherein a second frequency band comprises atleast the second transmit frequency band and the second receivefrequency band.
 7. The fixed wireless access radio of claim 6, whereinthe second frequency band is either within a frequency range of between2 GHz and 7 GHz or within a frequency range of above 10 GHz.
 8. Thefixed wireless access radio of claim 1, wherein at least one antennaelement of the plurality of antenna elements is a directive gain antennaelement.
 9. The fixed wireless access radio of claim 1, wherein at leastone antenna element of the plurality of antenna elements is at least oneof a patch antenna element, a dipole antenna element, or a slot antennaelement.
 10. The fixed wireless access radio of claim 1, wherein atleast one antenna element of the plurality of antenna elements iscoupled or couplable to at least one receive RF chain or transmit RFchain via either at least one RF switch or at least one duplexer filter.11. The fixed wireless access radio of claim 1, wherein the fixedwireless access radio is configured to use at least a Single-CarrierFrequency Domain Equalization (SC-FDE) modulation format.
 12. The fixedwireless access radio of claim 1, wherein the fixed wireless accessradio is configured to use at least an Orthogonal Frequency DivisionMultiplexing (OFDM) modulation format.
 13. The fixed wireless accessradio of claim 1, wherein the fixed wireless access radio is configuredto transmit in the first transmit frequency band and receive in thefirst receive frequency band coincident in time for at least a period oftime.
 14. The fixed wireless access radio of claim 1, wherein the fixedwireless access radio is configured to transmit in the second transmitfrequency band and receive in the second receive frequency bandcoincident in time for at least a period of time.
 15. The fixed wirelessaccess radio of claim 1, wherein the surge throughput capability ishigher than the base throughput capability.
 16. The fixed wirelessaccess radio of claim 1, wherein the risk of temporal interferenceoutage for the surge throughput capability is higher than for the basethroughput capability.
 17. The fixed wireless access radio of claim 1,wherein at least one of the plurality of receive RF chains comprises atleast a vector demodulator and two analog to digital converters that areconfigured to produce a respective one of a plurality of receive chainoutput signals comprised of digital baseband quadrature signals.
 18. Thefixed wireless access radio of claim 1, wherein at least one of theplurality of transmit RF chains comprises at least a vector modulatorand two digital to analog converters that are configured to produce arespective one of the plurality of transmit RF signals from a respectiveone of a plurality of transmit chain input signals comprised of digitalbaseband quadrature signals.
 19. A fixed wireless access radio forexchanging one or more data interface streams with one or more otherfixed wireless access radios, said fixed wireless access radiocomprising: a plurality of receive radio frequency (RF) chains, whereineach of the plurality of receive RF chains is configured to convert froma respective one of a plurality of receive RF signals within a receivefrequency band to a respective one of a plurality of receive chainoutput signals; a plurality of transmit radio frequency (RF) chains,wherein each of the plurality of transmit RF chains is configured toconvert from a respective one of a plurality of transmit chain inputsignals to a respective one of a plurality of transmit RF signals withina transmit frequency band; a plurality of directive gain antennaelements, wherein each of the plurality of directive gain antennaelements is configured to operate over at least both of the transmitfrequency band and the receive frequency band; and a plurality ofduplexer filters, wherein each duplexer filter comprises at least areceive band-select filter configured to selectively pass RF signalswithin the receive frequency band and a transmit band-select filterconfigured to selectively pass RF signals within the transmit frequencyband, wherein each duplexer filter is couplable or coupled to at leastone of the plurality of directive gain antenna elements, wherein thereceive band-select filter of each duplexer filter is couplable orcoupled to at least one of the plurality of receive RF chains, andwherein the transmit band-select filter of each duplexer filter iscouplable or coupled to at least one of the plurality of transmit RFchains; wherein the fixed wireless access radio is configured to operateat least a first subset of the plurality of transmit RF chains at afirst transmit RF carrier frequency and to operate at least a secondsubset of the plurality of transmit RF chains at a second transmit RFcarrier frequency; and wherein the fixed wireless access radio isfurther configured to select at least one of the first transmit RFcarrier frequency or the second transmit RF carrier frequency inresponse to at least a current link condition at an at least one of theone or more other fixed wireless access radios.
 20. The fixed wirelessaccess radio of claim 19, further comprising: one or more demodulatorcores, wherein each demodulator core is configured to demodulate one ormore of a plurality of receive symbol streams to produce one or morereceive data interface streams; and a frequency selective receive pathchannel multiplexer, interposed between the one or more demodulatorcores and at least the plurality of receive RF chains, wherein thefrequency selective receive path channel multiplexer is configured togenerate the plurality of receive symbol streams from at least theplurality of receive chain output signals.
 21. The fixed wireless accessradio of claim 20, wherein each one of the one or more demodulator corescomprises at least a decoder and a soft decision symbol demapper; andwherein each one of the plurality of receive RF chains comprises atleast a vector demodulator and two analog to digital converters that areconfigured to produce the respective one of the plurality of receivechain output signals, each said respective one of the plurality ofreceive chain output signals comprised of digital baseband quadraturesignals.
 22. The fixed wireless access radio of claim 19, wherein eachone of the plurality of transmit RF chains comprises at least a vectormodulator and two digital to analog converters that are configured toproduce the respective one of the plurality of transmit RF signals, eachsaid respective one of the plurality of transmit chain input signalscomprised of digital baseband quadrature signals.
 23. The fixed wirelessaccess radio of claim 20, wherein each one of the one or moredemodulator cores comprises at least one of a descrambler or adeinterleaver; and wherein each one of the one or more modulator corescomprises at least one of a scrambler or an interleaver.
 24. The fixedwireless access radio of claim 19, further comprising: one or moreselectable RF connections that are configured to selectively couplecertain of the plurality of directive gain antenna elements to either orboth of certain of the plurality of receive RF chains or certain of theplurality of transmit RF chains; wherein the number of directive gainantenna elements that are configured to be selectively coupled toreceive RF chains exceeds the number of receive RF chains that areconfigured to accept receive RF signals from the one or more selectableRF connections; or wherein the number of directive gain antenna elementsthat are configured to be selectively coupled to transmit RF chainsexceeds the number of transmit RF chains that are configured to providetransmit RF signals to the one or more selectable RF connections. 25.The fixed wireless access radio of claim 24, wherein at least one of theone or more selectable RF connections comprises at least one RF switch.26. The fixed wireless access radio of claim 24, wherein the set ofreceive RF chains that is configured to accept receive RF signals fromthe one or more selectable RF connections is divided between a firstsubset that is configured to accept receive RF signals from directivegain antenna elements with a first polarization and a second subset thatis configured to accept receive RF signals from directive gain antennaelements with a second polarization; or wherein the set of transmit RFchains that is configured to provide transmit RF signals to the one ormore selectable RF connections is divided between a third subset that isconfigured to provide transmit RF signals to directive gain antennaelements with a first polarization and a fourth subset that isconfigured to provide transmit RF signals to directive gain antennaelements with a second polarization.
 27. The fixed wireless access radioof claim 19, wherein the directive gain antenna elements are arranged ona plurality of facets with one or more directive gain antenna elementsper facet, and wherein each facet is oriented at a different azimuthangle relative to at least one other facet.
 28. The fixed wirelessaccess radio of claim 19, further comprising: a plurality of poweramplifiers, wherein each power amplifier is configured to amplify atleast one of the transmit RF signals, and wherein each power amplifieris couplable or coupled to at least one of the plurality of transmit RFchains and to at least one transmit band-select filter of the pluralityof duplexer filters; and a plurality of low noise amplifiers, whereineach low noise amplifier is configured to amplify at least one of thereceive RF signals, and wherein each low noise amplifier is couplable orcoupled to at least one of the plurality of receive RF chains and to atleast one receive band-select filter of the plurality of duplexerfilters.
 29. The fixed wireless access radio of claim 19, wherein thefirst transmit frequency band is coincident with the first receivefrequency band.
 30. The fixed wireless access radio of claim 19, whereinfirst frequency band comprises at least the first transmit frequencyband and the first receive frequency band.
 31. The fixed wireless accessradio of claim 19, wherein the first frequency band is either within afrequency range of between 2 GHz and 7 GHz or within a frequency rangeof above 10 GHz.
 32. The fixed wireless access radio of claim 19,wherein the second transmit frequency band is coincident with the secondreceive frequency band.
 33. The fixed wireless access radio of claim 19,wherein second frequency band comprises at least the second transmitfrequency band and the second receive frequency band.
 34. The fixedwireless access radio of claim 19, wherein the second frequency band iseither within a frequency range of between 2 GHz and 7 GHz or within afrequency range of above 10 GHz.
 35. The fixed wireless access radio ofclaim 20, wherein the frequency selective receive path channelmultiplexer comprises at least one of a Space Division Multiple Access(SDMA) combiner or equalizer, a maximal ratio combining (MRC) combineror equalizer, a minimum mean squared error (MMSE) combiner or equalizer,an Eigen Beam Forming (EBF) combiner or equalizer, a receive beamforming (BF) combiner or equalizer, a Zero Forcing (ZF) combiner orequalizer, a channel estimator, a Maximal Likelihood (DL) detector, anInterference Canceller (IC), a VBLAST combiner or equalizer, a DiscreteFourier Transformer (DFT), a Fast Fourier Transformer (FFT), or anInverse Fast Fourier Transformer (IFFT).
 36. The fixed wireless accessradio of claim 20, wherein the frequency selective receive path channelmultiplexer comprises: a plurality of cyclic prefix removers, whereineach cyclic prefix remover is configured to discard a fraction of anoverall number of samples within one or more blocks of a plurality ofblocks of samples from a respective one of the plurality of receivechain output signals to produce a respective cyclic prefix removed oneor more blocks of samples, said fraction corresponding to a known cyclicprefix length for a plurality of second transmit symbol streams expectedto be comprised within the plurality of receive chain output signals; aplurality of respective complex Discrete Fourier Transformers coupled toeach respective cyclic prefix remover, wherein each complex DiscreteFourier Transformer is configured to decompose the respective cyclicprefix removed one or more blocks of samples into a respective set ofreceive chain frequency domain subchannel samples; and a plurality ofreceive channel equalizers coupled to the plurality of respectivecomplex Discrete Fourier Transformers, wherein each receive channelequalizer is configured to produce a set of channel-equalized frequencydomain estimates representative of a respective one of the plurality ofsecond transmit symbol streams by applying respective stream-specificand chain-specific receive weights to the respective sets of receivechain frequency domain subchannel samples; wherein said respectivestream-specific and chain-specific receive weights applied to therespective sets of receive chain frequency domain subchannel samplesvary with relative frequency domain subchannel position within suchsets.
 37. The fixed wireless access radio of claim 36, furthercomprising: a channel equalizer coefficients generator, wherein thechannel equalizer coefficients generator is configured to determine therespective stream-specific and chain-specific receive weights based atleast upon comparison of certain sets of receive chain frequency domainsubchannel samples with certain expected blocks of known frequencydomain subchannel samples expected to be present at certain times withinthe plurality of receive chain output signals.
 38. The fixed wirelessaccess radio of claim 36, further comprising: a plurality of complexInverse Discrete Fourier Transformers, wherein each complex InverseDiscrete Fourier Transformer is configured to compose a respective oneof the plurality of receive symbol streams from respective sets ofchannel-equalized frequency domain estimates representative of therespective one of the plurality of second transmit symbol streams. 39.The fixed wireless access radio of claim 38, wherein each of theplurality of complex Inverse Discrete Fourier Transformers isimplemented by a structure executing a complex Inverse Fast FourierTransform (IFFT), and wherein each of the plurality of complex DiscreteFourier Transformers is implemented by a structure executing a complexFast Fourier Transform (FFT).
 40. The fixed wireless access radio ofclaim 36, wherein each of the plurality of receive channel equalizerscomprises a number of complex multipliers corresponding to a number ofthe plurality of receive chain output signals, and a combiner.
 41. Thefixed wireless access radio of claim 36, wherein a receive pathmodulation format is based upon Orthogonal Frequency DivisionMultiplexing (OFDM).
 42. The fixed wireless access radio of claim 38,wherein a receive path modulation format is based upon Single-CarrierFrequency Domain Equalization (SC-FDE).
 43. The fixed wireless accessradio of claim 19, further comprising: one or more modulator cores,wherein each modulator core is configured to modulate one or moretransmit data interface streams to produce one or more of a plurality oftransmit symbol streams, wherein each transmit symbol stream comprisesat least a plurality of blocks of symbols, and wherein each one of theone or more modulator cores comprises at least an encoder and a symbolmapper; a non-frequency selective transmit path channel multiplexer,interposed between the one or more modulator cores and at least theplurality of transmit RF chains, wherein the non-frequency selectivetransmit path channel multiplexer is configured to generate theplurality of transmit chain input signals from at least the plurality oftransmit symbol streams; wherein a first number of the plurality oftransmit chain input signals exceeds a second number of the plurality oftransmit symbol streams; and wherein the non-frequency selectivetransmit path channel multiplexer is configured to apply respective setsof stream-specific and chain-specific transmit beamforming weights to atleast one or more blocks of the plurality of blocks of symbols from theplurality of transmit symbol streams when generating a respective one ofthe plurality of transmit chain input signals, and wherein a particularone of said stream-specific and chain-specific transmit beamformingweights is invariant with respect to a relative symbol position withinsaid at least one or more blocks of the plurality of blocks of symbols.44. The fixed wireless access radio of claim 43, wherein thenon-frequency selective transmit path channel multiplexer comprises: aplurality of cyclic prefix adders, wherein each cyclic prefix adder isconfigured to add a fraction of an overall number of samples within oneor more blocks of a plurality of blocks of samples corresponding to arespective one of the plurality of transmit chain input signals, saidfraction corresponding to a pre-determined cyclic prefix length; and aplurality of transmit channel equalizers, wherein each transmit channelequalizer is configured to produce one or more blocks of non-frequencyselective, channel-equalized samples corresponding to a respective oneof the plurality of transmit chain input signals by applying respectivesets of the stream-specific and chain-specific transmit beamformingweights to corresponding blocks of symbols from the plurality oftransmit symbol streams; wherein a number of the plurality of cyclicprefix adders and of the plurality of transmit channel equalizerscorresponds to the first number.
 45. The fixed wireless access radio ofclaim 44, further comprising: a plurality of complex Inverse DiscreteFourier Transformers, wherein each complex Inverse Discrete FourierTransformer is configured to compose a respective one of the pluralityof transmit chain input signals from respective ones of non-frequencyselective, channel-equalized samples corresponding to respective ones ofthe plurality of transmit chain input signals.
 46. The fixed wirelessaccess radio of claim 44, wherein an output from each respective one ofthe plurality of transmit channel equalizers is coupled to an input of arespective one of the plurality of cyclic prefix adders.
 47. The fixedwireless access radio of claim 44, wherein each of the plurality oftransmit channel equalizers comprises a number of complex multiplierscorresponding to the second number, and a combiner.
 48. The fixedwireless access radio of claim 44, wherein the stream-specific andchain-specific transmit beamforming weights are determined at a receivercomprised within at least one of the fixed wireless access radio or theone or more other fixed wireless access radios.
 49. The fixed wirelessaccess radio of claim 48, wherein the receiver that determines thestream-specific and chain-specific transmit beamforming weights furthercomprises: a channel equalizer coefficients generator, wherein thechannel equalizer coefficients generator is configured to determine therespective stream-specific and chain-specific transmit beamformingweights based at least upon comparison of certain signals at thereceiver with certain expected signals expected to be present at certaintimes.
 50. The fixed wireless access radio of claim 44, wherein thestream-specific and chain-specific transmit beamforming weights aredetermined in order to improve either a signal to interference and noiseratio (SINR) or a signal to noise ratio (SNR).
 51. The fixed wirelessaccess radio of claim 47, wherein each of the stream-specific andchain-specific transmit beamforming weights comprises at least a realbranch component and an imaginary branch component.
 52. The fixedwireless access radio of claim 44, wherein each of the stream-specificand chain-specific transmit beamforming weights comprises at least oneof an amplitude component or a phase component.
 53. The fixed wirelessaccess radio of claim 44, wherein a transmit path modulation format isbased upon Single-Carrier Frequency Domain Equalization (SC-FDE). 54.The fixed wireless access radio of claim 45, wherein a transmit pathmodulation format is based upon Orthogonal Frequency DivisionMultiplexing (OFDM).
 55. The fixed wireless access radio of claim 19,further comprising: a radio resource controller (RRC); wherein the radioresource controller is configured to select the at least one of thefirst transmit RF carrier frequency or the second transmit RF carrierfrequency in response to at least the current link condition at the atleast one of the one or more other fixed wireless access radios.
 56. Thefixed wireless access radio of claim 19, wherein a first channelbandwidth corresponding to the first transmit RF carrier frequency isequal to a second channel bandwidth corresponding to the second transmitRF carrier frequency.
 57. The fixed wireless access radio of claim 19,wherein a first channel bandwidth corresponding to the first transmit RFcarrier frequency is not equal to a second channel bandwidthcorresponding to the second transmit RF carrier frequency.
 58. The fixedwireless access radio of claim 19, wherein the current link condition isderived from at least one or more link quality metrics determined at theat least one of the one or more other fixed wireless access radios. 59.The fixed wireless access radio of claim 58, wherein the at least one ormore link quality metrics comprise at least one or more of a receivestrength signal indication (RSSI), a decoder metric, a frame check sum(FCS) failure rate, a signal to noise ratio (SNR) or a signal tointerference and noise ratio (SINR).
 60. The fixed wireless access radioof claim 19, wherein the fixed wireless access radio is configured tooperate at least a first subset of the plurality of receive RF chains ata first receive RF carrier frequency and to operate at least a secondsubset of the plurality of receive RF chains at a second receive RFcarrier frequency.
 61. A fixed wireless access radio for exchanging oneor more data interface streams with one or more other fixed wirelessaccess radios, said fixed wireless access radio comprising: a pluralityof receive radio frequency (RF) chains, wherein each of the plurality ofreceive RF chains is configured to convert from a respective one of aplurality of receive RF signals within a receive frequency band to arespective one of a plurality of receive chain output signals; aplurality of transmit radio frequency (RF) chains, wherein each of theplurality of transmit RF chains is configured to convert from arespective one of a plurality of transmit chain input signals to arespective one of a plurality of transmit RF signals within a transmitfrequency band; and a plurality of directive gain antenna elements,wherein a first subset of the plurality of directive gain antennaelements is configured to operate over at least the transmit frequencyband and is couplable or coupled to at least one the plurality oftransmit RF chains and a second subset of the plurality of directivegain antenna elements is configured to operate over at least the receivefrequency band and is couplable or coupled to at least one the pluralityof receive RF chains; wherein the fixed wireless access radio isconfigured to operate at least a first subset of the plurality oftransmit RF chains at a first transmit RF carrier frequency and tooperate at least a second subset of the plurality of transmit RF chainsat a second transmit RF carrier frequency; and wherein the fixedwireless access radio is further configured to select at least one ofthe first transmit RF carrier frequency or the second transmit RFcarrier frequency in response to at least a current link condition at anat least one of the one or more other fixed wireless access radios. 62.The fixed wireless access radio of claim 61, wherein each of theplurality of directive gain antenna elements is configured to operateover at least both of the transmit frequency band and the receivefrequency band.
 63. The fixed wireless access radio of claim 62, whereinthe transmit frequency band is coincident with the receive frequencyband.